Semiconductor device and power supply device

ABSTRACT

A power supply topology is used in which a transistor is provided on the side of an output node of a rectifying circuit. An inductor is provided on the side of a reference node, a resistor is inserted between the transistor and the inductor, and one end of the resistor is coupled to a ground power supply voltage of a PFC circuit. The PFC circuit includes a square circuit which squares a result of multiplication of an input voltage detection signal and feedback information (output voltage of an error amplifier circuit). The PFC circuit drives on the transistor when a detection voltage developed at the resistor reaches zero, and drives off the transistor when the detection signal reaches an output signal of the square circuit.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.14/695,517, filed Apr. 24, 2015, which is a continuation of U.S. patentapplication Ser. No. 14/272,701, filed May 8, 2014, now U.S. Pat. No.9,041,314, which is a continuation of U.S. patent application Ser. No.13/109,423, filed May 17, 2011, now U.S. Pat. No. 8,754,590, whichclaims priority to Japanese Patent Application No. 2010-122971, filedMay 28, 2010, and Japanese Patent Application No. 2010-257090, filedNov. 17, 2010, the disclosures of which, including the specifications,drawings and abstracts, are incorporated herein by reference in theirentirety.

BACKGROUND

The present invention relates to a semiconductor device and a powersupply device using the same, and in particular to a technologyeffective when applied to a switching power supply device equipped witha power factor correction circuit.

A technology which controls a pulse width of a switch using a squarecircuit and an adder to reduce a distortion rate of an input current ina switching power supply device based on an input electrolyticcapacitor-less one converter system, has been described in, for example,a patent document 1.

-   [Patent document 1] Japanese Unexamined Patent Publication No.    2002-300780.

SUMMARY

Energy saving has recently been accelerated in various fields. As one ofthem, an LED bulb using light emitting diodes (LED) good in lightemission efficiency is rapidly becoming pervasive in, for example, anillumination field instead of a filament lamp. The LED bulb is driven byapplying power generated by, for example, an AC-DC converter or the liketo each of light emitting diodes. FIG. 28 is a schematic diagramillustrating a configuration example of an AC-DC converter discussed asthe premise of the present invention. FIG. 29 is a waveform diagramshowing an example of operation of FIG. 28. The AC-DC converter shown inFIG. 28 serves as a step-up power converter (step-up converter)including a PFC (PFC: power factor correction) circuit. The AC-DCconverter using a commercial power source is generally equipped with aPFC circuit to avoid the occurrence of obstacles (malfunction,generation of heat, burnout, etc.) due to harmonic current in otherdevices via an AC power line.

In FIG. 28, a commercial power source (AC) (e.g., 85 to 264 Vrms or thelike) is first full-wave rectified by a rectifying circuit DB1. Inputpower from the rectifying circuit DB1 is stored or accumulated in aninductor LM1 of a transformer TR2 when a transistor Q1 is on, and thepower stored in the inductor LM1 is discharged to an output capacitorCout via a diode D1 when the transistor Q1 is off. At this time, the PFCcircuit PIC10 is inputted with information (Vz) about an input currentIin flowing into the inductor LM1, detected via a supplementary windingLMs of the transformer TR2, information (Vin′) about an input voltageVin from the rectifying circuit DB1, information (Vout′) about an outputvoltage Vout, and information (Vcs) about a current Iq1 flowing throughthe transistor Q1. The PFC circuit PIC10 drives on the transistor Q1(i.e., operates in a current critical mode) when it has detected thatthe input current Iin has reached zero due to the information Vz. ThePFC circuit PIC10 drives the transistor Q1 off when it has detected thatthe current Iq1 has reached the predetermined times (multiple numbercorresponding to Vout′) the Vin′ due to the information Vcs.Consequently, the input current Iin (ac current Iac flowing through ACpower line) flowing through the inductor LM1 becomes a sinusoidal shape,thus enabling a reduction in harmonic current developed in the AC powerline.

Such an AC-DC converter as shown in FIG. 28 is however subject to thelimitation that the output voltage Vout should be set higher than theinput voltage Vin (e.g., 85 to 264 Vrms) (e.g., Vout=390 V or the likein the case of worldwide adaptation). Therefore, circuit parts such asthe output capacitor Cout having stored the information Vout therein, acontrol circuit coupled to a subsequent stage, etc. need to use partshaving a high breakdown voltage (e.g., 400 V or higher). It is fearedthat an increase in part cost and a size increase in power supply devicewill occur. Further, when a device that needs not have a high voltage sofar is driven as typified by an illumination field, for example, thereis a need to provide a step-down power converter (step-down converter)DWC in a stage subsequent to such a step-up converter UPC as shown inFIG. 28 as shown in a power supply system of FIG. 30, for example. Inthis case, a further increase in part cost and a further size increasein the power supply system are feared.

Thus, in order to solve such a problem, for example, such a flyback typeAC-DC converter as shown in FIG. 31 may be used in the illuminationfield or the like. FIG. 31 is a schematic diagram illustrating anotherconfiguration example of the AC-DC converter discussed as the premise ofthe present invention. In FIG. 31, in a manner similar to FIG. 28, inputpower from a rectifying circuit DB1 is first stored or accumulated in atransformer TR1 via a primary winding (inductor) LM1 of the transformerTR1 when a transistor Q12 is on. Unlike the case of FIG. 28, however,the power stored in the transformer TR1 is discharged from a secondarywinding (inductor) LM2 of the transformer TR1 to an output capacitorCout via a diode D1 when the transistor Q1 is off.

A PFC circuit PIC10 of FIG. 31 has a configuration similar to that ofFIG. 28 and is inputted with information (Vz) about the power stored inthe transformer TR1 detected via its supplementary winding LMs of thetransformer TR1, information (Vin′) about an input voltage Vin from therectifying circuit DB1, information (Vout′) about an output voltageVout, and information (Vcs) about an input current Iin flowing throughthe transistor Q12. When the stored power of the transformer TR1 hasreached zero (in other words, the output current lout flowing throughthe inductor LM2 becomes zero), the PFC circuit PIC10 drives on thetransistor Q1 (i.e., it operates in a current critical mode). When theinput current Iin flowing through the transistor Q1 via the Vcs hasreached the predetermined times (multiple number corresponding to Vout′)the Vin′, the PFC circuit PIC10 drives off the transistor Q1. When sucha configuration example is used, for example, the number of turns n1 inthe primary winding LM1 of the transformer TR1 is set greater than thenumber of turns n2 in the secondary winding LM2 when a device that needsnot have a high voltage so far is driven, as typified by an illuminationfield, and the example of FIG. 31 may be operated as the step-downconverter DWC. Consequently, as shown in a power supply system of FIG.33, a reduction in the number of parts is enabled and a reduction incost, a size reduction in the power supply system, etc. can be achievedas compared with the case of FIG. 30.

It is however feared that the following problems will occur where such aconfiguration example as shown in FIG. 31 is used. The first problem isthat the waveform of an ac current Iac flowing through an AC power linebecomes a shape distorted like a trapezoid as shown in FIG. 34(b). FIGS.34(a) and 34(b) respectively show one example of the waveform of accurrent flowing through the AC power line, in which FIG. 34(a) is awaveform diagram where the step-up converter of FIG. 28 is used, andFIG. 34(b) is a waveform diagram where the flyback converter of FIG. 31is used. Thus, the ac current waveform Iac becomes a sinusoidal wavewhen the configuration example of FIG. 28 is used, whereas when theconfiguration example of FIG. 31 is used, the ac current waveform Iacbecomes a distorted waveform. This distortion reduces a power factor andcauses harmonic current on the AC power line.

The difference between FIGS. 34(a) and 34(b) qualitatively results fromthe fact that when the configuration example of FIG. 28 is used, theinput current Iin flows continuously during an on/off period of thetransistor Q1 as shown in FIG. 29, whereas when the configurationexample of FIG. 31 is used, the input current Iin flows only during theon period of the transistor Q1 as shown in FIG. 32. Described in moredetails, this is because when the configuration example of FIG. 28 isused, the input current Iin becomes a value proportional to the inputvoltage Vin indicative of the sinusoidal wave as expressed in thefollowing equation (1A), whereas when the configuration example of FIG.31 is used, a simple proportional relation is not established betweenthe Iin and the Vin as expressed in the following equation (2A).

$\begin{matrix}\begin{matrix}{{I\; {in}} = {{\frac{K}{2 \times R\; {cs}} \cdot V}\; {{in}\left( {K = {{circuit}\mspace{14mu} {design}\mspace{14mu} {constant}}} \right)}}} & \;\end{matrix} & \left( {1A} \right) \\{{I\; {in}} = {\frac{K}{2 \times R\; {cs}} \cdot \frac{V\; {in} \times {\left( {n\; {1/n}\; 2} \right) \cdot V}\; {out}}{{V\; {in}} + {{\left( {n\; {1/n}\; 2} \right) \cdot V}\; {out}}}}} & \left( {2A} \right)\end{matrix}$

(K: circuit design constant)

As the second problem, there may be mentioned an increase in part costand a size increase in power supply device. Although the use of theconfiguration example of FIG. 31 enables the reduction in part cost andthe size reduction in the power supply device as compared with theconfiguration example of FIG. 28, there is a demand for furtherminiaturization, a further reduction in part cost and the like where theconfiguration example of FIG. 31 is applied to, for example, an LED bulbor the like. In the configuration example of FIG. 31, the size of thetransformer TR1 equipped with the primary winding LM1, secondary windingLM2 and supplementary winding LMs is large in particular. When theinductance value of the primary winding LM1 is set to 1 mH or the like,for example, the transformer TR1 can take sizes like the height, widthand breadth that can be rendered to be 15 mm or so respectively.

As the third problem, there is mentioned a reduction in power conversionefficiency. In such a flyback system as shown in FIG. 31, a relativelylarge loss occurs due to a current (flyback current) flowing into theprimary winding LM1 side when the power is taken out from the secondarywinding LM2 side of the transformer TR1 at the time of off of thetransistor Q1. In general, for example, the power conversion efficiencyof such a step-up converter as shown in FIG. 28 is 95% or the like,whereas the power conversion efficiency of such as flyback converter asshown in FIG. 31 is 85% or the like.

The present invention has been made in view of the foregoing. The aboveand other objects and novel features of the present invention will beapparent from the description of the specification and the accompanyingdrawings.

A summary of typical embodiments of the invention disclosed in thisapplication will be briefly described as follows:

A power supply device according to the present embodiment includes arectifying circuit, a switch element, an inductor, a current detectionresistor, a control circuit, an input voltage detection circuit, a diodeand an output capacitor. The power supply device is capable of improvinga power factor of a current waveform flowing into the rectifying circuitby on/off-control of the switch element by the control circuit. Therectifying circuit rectifies ac power and supplies power to a first nodeon the basis of a second node. The switch element has one end coupled tothe first node. The inductor has one end coupled to the second node. Theinductor stores the power of the first node therein via the switchelement when the switch element is driven on, and discharges the storedpower when the switch element is driven off. The current detectionresistor is inserted between the other end of the switch element and theother end of the inductor. The control circuit allows a third node usedas one end of the current detection resistor to function as a groundpower supply voltage and controls on/off of the switch element. Theoutput capacitor and diode are inserted onto a path for discharging thepower stored in the inductor. The input voltage detection circuitoutputs a first voltage by resistance division between the first nodeand the third node. Here, the control circuit drives off the switchelement using a second voltage developed at a fourth node used as theother end of the current detection resistor, and the first voltage tothereby perform an improvement in power factor.

Improving the power factor using such a power supply topology enablesreductions in the size of a power supply device and the entire powersupply system including the power supply device. Upon performing acurrent critical mode operation in particular, a control circuit detectsa second voltage from a current detection resistor to thereby determinea timing provided to drive on a switch element and a timing provided todrive off the switch element, thus resulting in the usefulness in termsof a size reduction and the like.

A semiconductor device according to the present embodiment includesfirst through fourth terminals, a multiplication circuit, a squarecircuit, a first comparator circuit, and a second comparator circuit andassumes a function of improving a power factor. A first voltageproportional to a voltage obtained by rectifying ac power is inputted tothe first terminal. The second terminal outputs an on level for drivingon an external switch element and an off level for driving off theexternal switch element. When the second terminal outputs the on level,power is accumulated in an external inductor via the switch element.When the second terminal outputs the off level, the power of theinductor is discharged. A feedback signal outputted from an externalload driven by the power of the inductor is inputted to a thirdterminal. A second voltage proportional to a current flowing through theinductor is inputted to a fourth terminal. A multiplication circuitmultiplies the feedback signal and the first voltage by each other. Asquare circuit performs a square arithmetic operation on an outputvoltage of the multiplication circuit. A first comparator circuitdetects that the second voltage has reached an output voltage of thesquare circuit. A second comparator circuit detects that the currentflowing through the inductor has reached a zero level, based on thesecond voltage. Here, the second terminal outputs the on level inresponse to a detection signal from the second comparator circuit andoutputs the off level in response to a detection signal from the firstcomparator.

Thus, the timing provided to drive off a switch element based on asignal via a square circuit is controlled, thereby making it possible tobring an input current waveform closer to a sinusoidal wave bycombinations with several power supply topologies. Since the area is notso increased according to the provision of the square circuit, a sizereduction in a power supply device is also achieved along with animprovement in power factor. The semiconductor device is more useful inthe case of a combination with the above power supply topology of thepresent embodiment.

Advantageous effects obtained by the typical embodiments of theinvention disclosed in the present application will be brieflyexplained. A power supply device equipped with a power factor correctioncircuit can be rendered small-sized. It is also possible to achieve afurther improvement in power factor.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a first embodimentof the present invention;

FIG. 2 is a waveform diagram illustrating a schematic example ofoperation of FIG. 1;

FIG. 3 is an explanatory diagram showing the concept of operation of asquare circuit in FIG. 2;

FIG. 4 is a supplementary diagram of FIG. 1;

FIG. 5 is a waveform diagram showing a part of FIG. 2 in enlarged form;

FIG. 6 is a schematic diagram showing a circuit configuration of acomparative example of FIG. 1;

FIGS. 7(a) and 7(b) are respectively waveform diagrams showing resultsof simulation of input current waveforms where the power supply deviceshown in FIG. 1 is used;

FIG. 8 is an outline diagram illustrating a schematic configurationexample of an LED illuminating device equipped with the power supplydevice of FIG. 1;

FIG. 9 is a circuit diagram showing a detailed configuration example ofa multiplication circuit of the power supply device of FIG. 1;

FIG. 10 is a circuit diagram depicting a detailed configuration exampleof the square circuit of the power supply device of FIG. 1;

FIG. 11 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a second embodimentof the present invention;

FIG. 12 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a third embodimentof the present invention;

FIG. 13 is a waveform diagram showing a schematic example of operationof FIG. 12;

FIG. 14 is a waveform diagram illustrating a part of FIG. 13 in enlargedform;

FIGS. 15(a) and 15(b) are respectively waveform diagrams showing resultsof simulation of input current waveforms where the power supply deviceshown in FIG. 12 is used;

FIG. 16 is a schematic diagram showing one example of a modified circuitconfiguration of the power supply device of FIG. 12;

FIG. 17 is a schematic diagram illustrating one example of anothermodified circuit configuration of the power supply device of FIG. 12;

FIG. 18 is a schematic diagram showing one example of a further modifiedcircuit configuration of the power supply device of FIG. 12;

FIG. 19 is a schematic diagram depicting one example of a circuitconfiguration of a power supply device according to a fourth embodimentof the present invention;

FIG. 20 is a schematic diagram showing one example of a modified circuitconfiguration of the power supply device of FIG. 19;

FIG. 21 is a schematic diagram depicting one example of a circuitconfiguration of a power supply device according to a fifth embodimentof the present invention;

FIG. 22 is a schematic diagram showing one example of a modified circuitconfiguration of the power supply device of FIG. 21;

FIG. 23 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a sixth embodimentof the present invention;

FIG. 24 is a waveform diagram showing a schematic example of operationof FIG. 23;

FIGS. 25(a) and 25(b) show the details of an oscillator and a triangularwave generator in FIG. 23, in which FIG. 25(a) is a circuit diagramshowing a configuration example thereof, and FIG. 25(b) is a waveformdiagram showing an example of operation of FIG. 25(a);

FIG. 26 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a seventh embodimentof the present invention;

FIG. 27 is a waveform diagram showing a schematic example of operationof FIG. 26;

FIG. 28 is a schematic diagram illustrating a configuration example ofan AC-DC converter discussed as the premise of the present invention;

FIG. 29 is a waveform diagram showing an example of operation of FIG.28;

FIG. 30 is a block diagram showing a configuration example of an overallpower supply system to which the AC-DC converter shown in FIG. 28 isapplied;

FIG. 31 is a schematic diagram illustrating another configurationexample of the AC-DC converter discussed as the premise of the presentinvention;

FIG. 32 is a waveform diagram showing an example of operation of FIG.31;

FIG. 33 is a block diagram illustrating a configuration example of anoverall power supply system to which the AC-DC converter shown in FIG.31 is applied;

FIGS. 34(a) and 34(b) respectively show one example illustrative of awaveform of an ac current flowing through an AC power line, in whichFIG. 34(a) is a waveform diagram where a step-up converter shown in FIG.28 is used, and FIG. 34(b) is a waveform diagram where a flybackconverter shown in FIG. 31 is used;

FIG. 35 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to an eighth embodimentof the present invention;

FIGS. 36(a) and 36(b) are respectively typical diagrams each showing oneexample of the conditions that switching losses occur, in which FIG.36(a) shows where the power supply device of FIG. 35 is used, and FIG.36(b) shows, as its comparative example, where the power supply deviceof FIG. 1 is used;

FIG. 37 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a ninth embodimentof the present invention;

FIG. 38 is waveform diagram showing a schematic example of operation ofthe power supply device of FIG. 37; and

FIG. 39 is a diagram showing a result of verification of higherharmonics contained in an input current that flows in the power supplydevice of FIG. 37.

DETAILED DESCRIPTION

Whenever circumstances require it for convenience in the followingembodiments, the subject matter will be described by being divided intoa plurality of sections or embodiments. However, unless otherwisespecified in particular, they are not irrelevant to one another. Onethereof has to do with modifications, details, supplementaryexplanations and the like of some or all of the other. When reference ismade to the number of elements or the like (including the number ofpieces, numerical values, quantity, range, etc.) in the followingembodiments, the number thereof is not limited to a specific number andmay be greater than or less than or equal to the specific number unlessotherwise specified in particular and definitely limited to the specificnumber in principle.

It is further needless to say that components (including element orfactor steps, etc.) employed in the following embodiments are not alwaysessential unless otherwise specified in particular and considered to bedefinitely essential in principle. Similarly, when reference is made tothe shapes, positional relations and the like of the components or thelike in the following embodiments, they will include ones substantiallyanalogous or similar to their shapes or the like unless otherwisespecified in particular and considered not to be definitely so inprinciple, etc. This is similarly applied even to the above-describednumerical values and range.

Circuit elements that configure respective functional blocks of theembodiments are not limited in particular, but formed over asemiconductor substrate like monocrystalline silicon by an IC technologyof known CMOS (complementary MOS transistors) or the like. Preferredembodiments of the present invention will hereinafter be described indetail based on the accompanying drawings. Incidentally, the samereference numerals are respectively attached to the same components ormembers in all the drawings for describing the embodiments in principle,and their repetitive description will be omitted.

First Embodiment

<<Overall Circuit Configuration of Power Supply Device [1]>>

FIG. 1 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a first embodimentof the present invention. The power supply device shown in FIG. 1includes a rectifying circuit DB1, a power factor correction (PFC)circuit (semiconductor device) PIC1, a transistor (switch element) Q1,resistors Rac1, Rac2 and Rcs, an inductor L1, a diode D1, an outputcapacitor Cout, and a power supply generator VCCGEN. The outputcapacitor Cout is coupled between a positive polarity output node Vout(+) and a negative polarity output node Vout (−) and supplies an outputvoltage Vout obtained between the positive polarity output node Vout (+)and the negative polarity output node Vout (−) as a power supply voltageof a load circuit LOD. Here, a plurality of light emitting diodes LED[1] through LED [n] are shown as one example of the load circuit LOD.The LEDs [1] through LED [n] are respectively coupled in series in orderwith the Vout (+) side as the anode and the Vout (−) side as thecathode.

The rectifying circuit DB1 full-wave rectifies a commercial power source(AC) (ac voltage Vac and ac current Iac) by four diodes Da1, Da2, Db1and Db2, for example, and outputs an input voltage Vin and an inputcurrent Iin to a rectified output node Ndb1 with the positive polarityoutput node Vout (+) as the reference. The resistors Rac1 and Rac2 arecoupled in series between the rectified output node Ndb1 and a groundpower supply voltage GND1 and outputs, from a common coupling nodethereof, an input voltage detection signal Vin′ that assumes a valuegenerated by resistance voltage division from Vin. The transistor (nchannel-type power MOS transistor herein) Q1 has a drain coupled to therectified output node Ndb1, a source coupled to a node Nsw, and a gatedriven by a duty control signal PWM generated from the PFC circuit PIC1.The resistor Rcs is coupled between the node Nsw and the ground powersupply voltage GND1, and the inductor L1 is coupled between the groundpower supply voltage GND1 and the positive polarity output node Vout(+). The diode D1 has an anode coupled to the negative polarity outputnode Vout (−) and a cathode coupled to the node Nsw.

The PFC circuit (semiconductor device) PIC1 is operated by the groundpower supply voltage GND1 and a power supply voltage VCC and outputs aduty control signal PWM in response to the input voltage detectionsignal Vin′ and a detection voltage Vcs obtained at the node Nsw by theresistor Rcs. The power supply voltage VCC is generated by the powersupply generator VCCGEN here. The power supply generator VCCGEN storespower of the rectified output node Ndb1 in a capacitor Cvcc via a diodeDvcc1 and a resistor Rvcc1 at power-on and thereby generates the powersupply voltage VCC. On the other hand, when the power is stable, thepower supply generator VCCGEN stores power of the positive polarityoutput node Vout (+) in the capacitor Cvcc via a diode Dvcc2 and aresistor Rvcc2 and thereby generates the power supply voltage VCC.

The PFC circuit PIC1 includes a square circuit SQ, comparator circuitsCMPp and CMPz, a set/reset latch circuit SRLT, a driver circuit DRV, abuffer circuit BF, a switch circuit SW1, a low-pass filter circuit LPF,and an error amplifier circuit EA. When a reset signal RT is outputtedfrom the comparator circuit CMPp, the set/reset latch circuit SRLTdrives the duty control signal PWM to an ‘L’ level (off level) throughthe driver circuit DRV. When a set signal ST is outputted from thecomparator circuit CMPz, the set/reset latch circuit SRLT drives theduty control signal PWM to an ‘H’ level (on level) through the drivercircuit DRV. The comparator circuit CMPz outputs the set signal ST whenthe detection voltage Vcs at the switch node Nsw becomes lower than adesired comparison voltage Vr1.

The buffer circuit BF buffers the detection voltage Vcs at a gain 1 andoutputs the same to the low-pass filter circuit LPF through the switchcircuit SW1. The switch circuit SW1 is turned on during a period inwhich the set/reset latch circuit SRLT drives the duty control signalPWM to the ‘L’ level (off level). The error amplifier circuit EAamplifies a difference between the output voltage of the low-pass filtercircuit LPF and a predetermined comparison voltage Vr2. Themultiplication circuit MUL multiplies the output voltage of the erroramplifier circuit EA and the input voltage detection signal Vin′ by eachother and generates an output signal Vm that assumes the result of itsmultiplication. The square circuit SQ squares the output signal Vm andgenerates an output signal Vs that assumes the result of its square. Thecomparator circuit CMPp outputs the reset signal RT when the detectionvoltage Vcs exceeds the output signal Vs.

The power supply device shown in FIG. 1 mainly has two features. Thefirst feature resides in that control by the PFC circuit PIC1 isperformed using the power supply topology of the high-side inverseconverter in which the transistor Q1 is provided on the side of the highpotential output node (Ndb1) of the rectifying circuit DB1, the inductorL1 is provided on the side of the low potential output node (Vout (+))of the rectifying circuit DB1, and the output capacitor Cout is providedbetween the node lying between the transistor Q1 and the inductor L1,and the node Vout (+). In particular at this time, the power supplydevice is characterized in that the resistor Rcs is provided between thetransistor Q1 and the inductor L1, and one end of the resistor Rcs iscoupled to the ground power supply voltage GND1 of the PFC circuit PIC1and that the input voltage detection signal Vin′ is obtained byresistance voltage division between the node (Ndb1) and the ground powersupply voltage GND1, The second feature resides in that the squarecircuit SQ is provided in a stage subsequent to the multiplicationcircuit MUL, and the on-level period of the duty control signal PWM isdetermined by the output signal Vs of the square circuit SQ. The detailsof these features will be explained as appropriate subsequently.

<<Overall Circuit Operation of Power Supply Device [1]>>

FIG. 2 is a waveform diagram showing a schematic example of operation ofFIG. 1. FIG. 3 is an explanatory diagram showing the concept ofoperation of the square circuit SQ in FIG. 1. Firstly, when a dutycontrol signal PWM is at an ‘H’ level (on level) as shown in FIG. 2, thetransistor Q1 is driven on, so that a growing input current Iin flowsinto the inductor L1 through the resistor Rcs, thus resulting in theaccumulation of power in the inductor L1. With an increase in the inputcurrent Iin, a detection voltage Vcs developed at one end (node Nsw) ofthe resistor Rcs also increases in growing form.

On the other hand, when the detection voltage Vcs reaches the voltagevalue of the output signal Vs of the square circuit SQ, a reset signalRT is generated from the comparator circuit CMPp to transition the dutycontrol signal PWM to an ‘L’ level (off level), so that the transistorQ1 is driven off. In doing so, an output current Iout flows through apath of the positive polarity output node Vout (+), negative polarityoutput node Vout (−), diode D1 and resistor Rcs with the poweraccumulated in the inductor L1 as an electromotive force. The loadcircuit LOD (LED [1] to LED [n]) is driven by the output current Iout.During a period in which the transistor Q1 is off, the current (outputcurrent Iout) flowing through the inductor L1 is reduced in decreasingform, and the detection voltage Vcs developed at one end (Nsw) of theresistor Rcs also decreases in like manner. When the detection voltageVcs falls below a comparison voltage Vr1 of the comparator circuit CMPz,a set signal ST is generated so that the transistor Q1 is driven onagain.

The comparison voltage Vr1 is set to a value close to zero like a few mVto a few hundred of mV, for example. With its setting, a set signal(zero current detection signal) ST is generated when the value of theoutput current Iout becomes substantially zero, so that the transistorQ1 is driven on. That is, the power supply device shown in FIG. 1operates in a current critical mode. In FIG. 1, the detection voltageVcs (i.e., the result of detection of the output current Iout) in theperiod of off of the transistor Q1 is inputted to the low-pass filtercircuit LPF through the switch circuit SW1. The error amplifier circuitEA detects a difference between the detected value of Iout and a setvalue (Vr2) of Iout determined in advance. The multiplication circuitMUL reflects the result of detection by the error amplifier circuit EAon the input voltage detection signal Vin′ to generate an output signalVm. Since the input voltage detection signal Vin′ becomes a waveformproportional to an input voltage Vin that assumes a sine wave (strictlythe absolute value waveform of sinusoidal wave), the output signal Vmassumes the sine wave (strictly the absolute value waveform ofsinusoidal wave), and the voltage amplitude thereof becomes a waveformthat varies according to the result of detection by the error amplifiercircuit EA.

Since the output signal Vs of the square circuit SQ is a value obtainedby squaring the output signal Vm, it becomes such a wave shape that thesinusoidal wave is pointed as shown in FIG. 2. Determining the period ofthe ‘H’ level (On of transistor Q1) of the duty control signal PWM,based on the output signal Vs having such a wave shape as describedabove makes it possible to bring the ac current Iac developed in thecommercial power line (AC) close to the sinusoidal wave qualitatively insuch a manner as shown in FIG. 3. If the square circuit SQ is notprovided in FIG. 3, the power supply device of FIG. 1 is of such acircuit system that the input current Iin (ac current Iac of commercialpower source (AC)) flows only during the period in which the transistorQ1 is on, in a manner similar to the flyback converter of FIG. 31referred to above. For this reason, in FIG. 3, the wave shape assumes atrapezoidal current waveform (Iac′) to which convex components are addedwith the sinusoidal wave as the reference in a manner similar to FIG.34(b).

Thus, in order to cancel out the convex components of the trapezoidalwave (Iac′), the square circuit SQ generates an output signal Vs havingcancave components with the sinusoidal wave as the reference, and thepower supply device of FIG. 1 generates a virtual current waveform(Iac″) (current waveform generated assuming that Iin (Iac) flows duringan on/off period of the transistor Q1) based on the output signal Vs.Then, actually, Iin (Iac) flows only during the on period of thetransistor Q1. For this reason, the convex components are added to theconcave components, thus resulting in the acquisition of a currentwaveform (Iac) close to the sinusoidal wave. Incidentally, the squarecircuit SQ is not necessarily required to take the square from thisconcept, but may take a multiplier factor having convex components takenonly to cancel out the concave components.

Although not limited in particular here, the numerical values of therespective circuits in the power supply device of FIG. 1 will beexplained by citing their concrete examples in the following manner. Theinput voltage Vin is a value obtained by full-wave rectifying acommercial power source (AC) (ac voltage Vac) like 85 to 264 Vrms or thelike. The resistance values of the resistors Rac1 and Rac2 are 3 MΩ and27 kΩ or the like respectively. In this case, the input voltagedetection signal Vin′ becomes a voltage value equal to about 1/100 ofthe input voltage Vin. The power supply voltage VCC of the PFC circuitPIC1 is 20 V or the like. The inductance value of the inductor L1 is 1mH or the like, and the capacitance value of the output capacitor Coutis 2 μF or the like. The output voltage Vout developed at the positivepolarity output node Vout (+) on the basis of the Vout (−) is 60 V orthe like assuming that twenty LEDs each having a forward voltage 3 V arecoupled. The resistance value of the resistor Rcs is 1 Ω or the like.Assuming that the power consumption of the load circuit (LED) is 10 W orthe like, the current value of the output current Iout becomes 0.17 A(=10/60). In this case, power consumed at the resistor Rcs becomes 29 mW(=0.17×0.17×1), so the effect of a loss by this power consumption onpower conversion efficiency of the entire power supply device is small.

The sinusoidal input voltage detection signal Vin′ such as shown in FIG.2 is strictly generated when the transistor Q1 is off. When thetransistor Q1 is on, the input voltage detection signal Vin′ is fixed toapproximately the level of the ground power supply voltage GND1. Thus,actually, since a sinusoidal wave corresponding to the input voltagedetection signal Vin′ at the time of turning off of the transistor Q1 isoutputted to the multiplication circuit MUL, such a holding circuit asshown in FIG. 4, for example, is inserted into an input part of themultiplication circuit MUL. The holding circuit shown in FIG. 4 isprovided with a diode D2 in which the input voltage detection signalVin′ is used as an anode input, and a capacitor Ch and a resistor Rh(low-pass filter circuit comprised of Ch and Rh) coupled in parallelbetween the cathode of the diode D2 and the ground power supply voltageGND1. Respective device constants are determined from a low-pass cut-offfrequency, AC-voltage detection resistors (Rac1, Rac2), the response ofthe capacitor Ch to AC-voltage detection resistors (Rac1, Rac2), andnoise resistance, etc. For example, the low-pass filter circuit has sucha frequency characteristic as to allow an AC frequency (e.g., 50 Hz or60 Hz) to pass therethrough and to sufficiently cut off a switchingfrequency (e.g., greater than or equal to 25 kHz). Using such a holdingcircuit makes it possible to average the input voltage detection signalVin′ at the time of turning-off of the transistor Q1 on a time sequencebasis and shape the voltage on the cathode of the diode D2 into thesinusoidal wave corresponding to the input voltage detection signalVin′.

<<Detailed Circuit Operation of Power Supply Device [1]>>

A detailed example of operation of a principle part in the power supplydevice shown in FIG. 1 will next be explained with reference to FIG. 5.FIG. 5 is a waveform diagram showing a part of FIG. 2 in enlarged formand shows a duty control signal (gate voltage of transistor Q1) PWM anda current IL1 flowing through the inductor L1, both of which areextracted in FIG. 2. In FIG. 2, for convenience of explanation, onecycle of the duty control signal PWM becomes long and the peak value ofthe current IL1 flowing through the inductor L1 for every cycle differsgreatly. In practice, however, one cycle of the duty control signal PWMis short and the peak value Ipk of the current IL1 is assumed to beapproximately constant within a given short period of time as shown inFIG. 5. Assuming that the inductance value of the inductor L1 is L1, theon period of the transistor Q1 is Ton, the off period thereof is Toff,(the voltage across the resistor Rcs) <<Vout, and (voltage drop in thetransistor Q1) <<Vout, the peak value Ipk of the current LI1 isexpressed in the following equation (1B):

Ipk=(Vin/L1)×Ton=(Vin/Li)×Toff   (1B)

From the above equation (1B), one cycle Tsw of the duty control signalPWM is expressed in the following equation (2B):

Tsw=Ton+Toff=((Vin+Vout)/Vout)×Ton   (2B)

Using these equations, an average value Iin_ave of the input current Iinin one cycle of the duty control signal PWM is defined by the followingequation (3B):

$\begin{matrix}{{I\; {in\_ ave}} = {\frac{\left( {1/2} \right) \times {Ipk} \times {Ton}}{Tsw} = {{1/\left( {{2 \cdot L}\; 1} \right)} \times \frac{{Vin} \times {Vout}}{\left. {{Vin} + {Vout}}\; \right|} \times {Ton}}}} & \left( {3\; B} \right)\end{matrix}$

Since the ground power supply voltage GND1 of the PFC circuit PIC1 iscoupled between the transistor Q1 and the inductor L1, the input voltagedetection signal Vin′ is expressed in the following equation (4B) andassumes a value proportional to (the voltages Vin+Vout).

Vin′=(Rac2/(Rac1+Rac2))×(Vin+Vout)   (4B)

When (Rac2/(Rac1+Rac2)) in the equation (4B) is assumed to be K2, andthe output signal of the error amplifier circuit EA is assumed to be K1,the output signal Vm of the multiplication circuit MUL becomesVm=K1·K2·(Vin+Vout). Accordingly, the output signal Vs of the squarecircuit SQ is expressed in the following equation (5B). In the equation(5B), K=(K1·K2)².

Vs=K·(Vin+Vout)²   (5B)

The comparator circuit CMPp sets Rcs×Ipk=equation (5B) using Ipk of theequation (1B) to control the transistor Q1 to off when the detectionvoltage Vcs based on the resistor Rcs reaches the output signal Vs. As aresult, the on period Ton is expressed in the following equation (6B):

Ton=((K·L1)/Rcs)×(Vin+Vout)² /Vin   (6B)

Substituting the equation (6B) into the equation (3B) yields thefollowing equation (7B). The average value Iin_ave (input current Iin)becomes a value proportional to (Vin+Vout), thus resulting in a waveformclose to the sinusoidal wave.

Iin_ave=(K/(2·Rcs))×Vout·(Vin+Vout)   (7B)

Thus, the power supply device (PFC circuit PIC1) shown in FIG. 1acquires the input voltage detection signal Vin′ in the form of afunction of (Vin+Vout) under the concept of the numerical expression,squares the same to generate the term of (Vin+Vout)²/Vin in the Ton asexpressed in the equation (6B) and cancels out the value (Vin+Vout) ofthe denominator in the equation (3B) by the Ton. Thus, the average valueIin₁₃ ave of the equation (3B) is brought close to the function of theinput voltage Vin corresponding to the sinusoidal wave.

On the other hand, in order to bring the average value Iin_ave to theperfect function of Vin, such a configuration example as shown in FIG.6, for example, is considered to be used. FIG. 6 is a schematic diagramshowing a circuit configuration illustrative of a comparative exampleshown in FIG. 1. The PFC circuit PIC2 shown in FIG. 6 is provided with aconstant generator VOGEN for generating a function of a constant (Vout),a two-input adder ADD, and a three-input multiplication circuit MUL′.The two-input adder ADD subtracts the output (K2·Vout) of the constantgenerator VOGEN from the input voltage detection signal Vin′ shown inthe equation (4B) and thereby outputs K2·Vin. The three-inputmultiplication circuit MUL′ multiplies the output of the two-input adderADD, the input voltage detection signal Vin′ and the output signal K1 ofthe error amplifier circuit EA by one another and thereby generates anoutput signal Vm′(=K1·(K2)²)·(Vin+Vout)). The term of (Vin+Vout) isgenerated in the Ton where the output signal Vm′ is used instead of theoutput signal Vs of FIG. 1, so that the value (Vin+Vout) of thedenominator in the equation (3B) is canceled out to make it possible tobring the average value Iin_ave to the perfect function of Vin.

When, however, such a configuration example as shown in FIG. 6 is used,the following points are feared, for example. Firstly, it is feared thatthere is a need to provide adjustment external terminals in order togenerate the optimum constant (K2·Vout) from the constant generatorVOGEN. That is, the PFC circuit PIC1 is achieved by a smallsemiconductor package having a small number of external pins (e.g., 6pins or 8 pins), for example, but additional external terminals arerequired to cope with Vout and K2 variable depending on the product tobe applied, and hence miniaturization of the power supply device becomesdifficult. Since the two-input adder ADD and the three-inputmultiplication circuit MUL′ also become necessary, a size increase inthe PFC circuit (power supply device) and an increase in cost are alsofeared with its need. Thus, when such a power supply device as shown inFIG. 1 is used, a substantially sinusoidal-shaped input current waveformwhich is not a perfect sinusoidal wave but has sufficient quantity inpractical use (is capable of reducing higher harmonic components asrequired and sufficiently), can be generated owing to its small-sizedand low-cost configuration.

FIGS. 7(a) and 7(b) are waveform diagrams showing results of simulationof an input current waveform where the power supply device of FIG. 1 isused. One example illustrative of an ideal input current waveform (idealsystem) indicative of being a sinusoidal wave and a theoretical inputcurrent waveform (square circuit control) calculated by the equation(7B) where the power supply device of FIG. 1 is used, is shown in FIG.7(a). One example indicative of a result of verification by a powersupply circuit simulator as targeted for the power supply device of FIG.1 is shown in FIG. 7(b). The theoretical input current waveform (squarecircuit control), as shown in FIG. 7(a), its wave shape itself becomes asinusoidal wave as is understood from the equation (7B), but offsetdistortion calculated by (K/(2·Rcs))·Vout² exists in a zero crossingpoint. However, in the circuit operation of the entire power supplydevice as shown in FIG. 7(b), the effect of distortion at the zero crossis of a level free of a problem in particular, and sufficient waveformquality can be ensured in terms of a practical use.

<<Application Example of Power Supply Device [1]>>

FIG. 8 is an outline diagram showing a schematic configuration exampleof an LED illuminating device equipped with the power supply device ofFIG. 1. The LED illuminating device SYS_LED shown in FIG. 8 serves aslight-bulb type LED illumination and is equipped thereinside with aplurality of light emitting diodes LEDs, and a wiring board BD whichsupplies power to the LEDs and corresponds to the power supply device ofFIG. 1. For example, a PFC circuit (semiconductor device) PIC1,resistors Rac and Rcs, a rectifying circuit DB1, an AC line filtercircuit FLT, a transistor Q1, a diode D1, an output capacitor Cout, aninductor L1, etc. that serve as independent parts (package parts)respectively are mounted over the wiring board BD. The diameter of thewiring board BD is about 4 cm or the like, for example. Since the LEDilluminating device or the like needs to achieve the power supply deviceusing such a small wiring board BD in particular, there is a demand forminiaturization of each individual part (package parts).

When such a flyback converter as shown in FIG. 31 is used for example,there is a fear that since the size of the transistor TR1 is large inparticular (when the inductance value on the primary winding side is 1mH or so, for example, its height, width and breadth are 15 mm or sorespectively), miniaturization thereof is not sufficiently achieved.Thus, the use of the power supply device of FIG. 1 makes it unnecessaryto provide a transformer equipped with a primary winding and a secondarywinding, and makes it possible to determine timing provided to performswitching between both on and off levels of the duty control signal PWMaccording to information from the current detecting resistor Rcs. Forthis reason, a transformer equipped with supplementary windings alsobecomes unnecessary. Accordingly, the inductor L1 may be providedinstead of the transformer, so that a size reduction in the power supplydevice is enabled and a reduction in part cost can also be achieved. Theinductor L1 whose inductance value is 1 mH or so can be achieved involume equal to about one-half that of the transformer TR1 in FIG. 31,for example. Incidentally, the AC line filter circuit FLT in FIG. 8 isof a filter circuit for eliminating higher harmonic components inputtedfrom the commercial power source (AC) and is placed between thecommercial power source (AC) and the rectifying circuit DB1 in FIG. 1.

<<Details of Multiplication Circuit and Square Circuit>>

FIG. 9 is a circuit diagram showing a detailed configuration example ofthe multiplication circuit MUL in the power supply device of FIG. 1. Themultiplication circuit MUL shown in FIG. 9 is equipped with npn typebipolar transistors QN1 through QN6, pnp type bipolar transistors QP1and QP2, and current sources IS1, IS2, ISb and IS3. The current sourceIS1 is a variable current source by which a current value I1 varies inproportion to an input voltage IN1 (Vin′ of FIG. 1). The current sourceIS2 is of a variable current source by which a current value I2 variesin proportion to an input voltage IN2 (output voltage of error amplifiercircuit EA in FIG. 1). A current value I3 of the current source IS3 isproportional to an internal constant, and a current value of the currentsource ISb is determined by a design constant. The multiplicationcircuit shown in FIG. 9 is generally called a translinear circuit.Attention is paid to a closed loop circuit comprised of the npn typebipolar transistors QN1 through QN6 in FIG. 9. Thus, the relationship of“VBE1+VBE2+VBE3=VBE4+VBE5+VBE6” is established with emitter-basevoltages of the npn type bipolar transistors QN1 through QN6 as VBE1through VBE6.

On the other hand, corrector currents (emitter currents) that flowthrough the npn type bipolar transistors QN1, QN2, QN3, QN4, QN5 and QN6are I1, I2, Iqn3, Io, I3 and Iqn3 respectively. Here, the emitter-basevoltage VBE of each bipolar transistor is expressed in VBE≈V_(T)·1n(I/Is) using the corrector current (emitter current) I, constant V_(T)and constant Is assuming that its base current is negligible.Accordingly, the relationship of “I1·I2·Iqn3=Io·I3·Iqn3” is establishedand “Io=((I1·I2)/I3)”. By converting the current Io to a voltage value,the output signal Vm of FIG. 1 indicative of the result ofmultiplication of the input voltages IN1 (Vin′ of FIG. 1) and IN2(output voltage of error amplifier circuit EA of FIG. 1) is obtained.

FIG. 10 is a circuit diagram showing a detailed configuration example ofa square circuit SQ in the power supply device of FIG. 1. The squarecircuit SQ shown in FIG. 10 is equipped with npn type bipolartransistors QN11 through QN14, pnp type bipolar transistors QP11 andQP12, and current sources IS10 and ISb and serves as a translinearcircuit in a manner similar to FIG. 9. The current source IS10 is of avariable current source by which a current value I10 varies inproportion to an input voltage IN3 (Vm of FIG. 1). A current value Ib ofthe current source ISb is determined by a design constant. Attention ispaid to a closed loop circuit comprised of the npn type bipolartransistors QN11 through QN14 in FIG. 10. Thus, the relationship of“VBE11+VBE12=VBE13+VBE14” is established with emitter-base voltages ofthe npn type bipolar transistors QN11 through QN14 as VBE11 throughVBE14. On the other hand, collector currents (emitter currents) thatflow through the npn type bipolar transistors QN11, QN12, QN13 and QN14are I10, I10, Ib, and Io respectively. Accordingly, the relationship of“I10·I10=Ib·Io” is established and “Io=((I10)²/Ib)”. By converting thecurrent Io to a voltage, the output signal Vm of FIG. 1 in which theinput voltage IN3 (Vm of FIG. 1) is squared is obtained.

In generally, the multiplication circuit MUL and the square circuit SQcan be achieved by various circuit systems as typified by an arithmeticcircuit using an operational amplifier, for example. The arithmeticcircuit using the operational amplifier is however high in accuracy butlarge in its circuit area and also becomes large in power consumption.To this end, such translinear circuits as shown in FIGS. 9 and 10 areused, so that the circuit area can be reduced and power consumption canalso be made relatively small. Incidentally, the translinear circuit canbe achieved using the bipolar transistors herein, but may also beachieved using a subthreshold region (region in which a source-draincurrent exponentially rises with respect to a gate-source voltage) ofeach MOS transistor, for example. In this case, a further reduction incircuit area and a reduction in power consumption can be achieved.

<<Advantages of Power Supply Device [1]>>

Thus, principal advantages obtained by using the power supply deviceaccording to the first embodiment are summarized as follows: As thefirst advantage, a size reduction in the power supply device or areduction in its cost can be achieved. This advantage is obtained by, asdescribed above, firstly using the power supply topology of thehigh-side inverse converter (might also be called polarity inversechopper system), optimizing the position of insertion of the currentdetecting resistor Rcs and the point of coupling of the PFC circuit PIC1to the ground power supply voltage GND1, and making the transformerunnecessary. This advantage is also obtained by using the square circuitSQ in the PFC circuit PIC1 instead of the use of the complicatedcircuits such as the adder ADD, three-input multiplication circuit MUL′and the like such as shown in FIG. 6.

As the second advantage, higher harmonic components likely to occur inthe input current can be reduced. This advantage is obtained by enablingthe input voltage detection signal Vin′ to be acquired as the functionof (Vin+Vout) by the use of the square circuit SQ and the combinationthereof with the power supply topology of the high-side inverseconverter in addition to it as described above. As the third advantage,power conversion efficiency can be improved. This advantage is obtainedfrom the use of the power supply topology of the high-side inverseconverter without using the power supply topology using the transformerlike such a flyback converter.

Incidentally, in the power supply device of FIG. 1, the input voltagedetection signal Vin′ is multiplied by the multiplication circuit MULand thereafter squared by the square circuit SQ, but the input voltagedetection signal Vin′ may be multiplied by the multiplication circuitMUL after having been squared by the square circuit SQ. In this case,however, it is feared that since the input voltage detection signal Vin′is squared and inputted to the multiplication circuit MUL, thedifference in level between the signals inputted to the multiplicationcircuit MUL where the input voltage detection signal Vin′ is large andsmall, becomes very large, thus causing a difficulty in ensuring asignal operation range inside the PFC circuit. From this point of view,such a coupling order as shown in FIG. 1 may preferably be used. Whensuch a configuration as described in the patent document 1 is used, forexample, the above coupling order is required to be set to the latterfor the reasons of the need for gain control of the square circuit. Inthe present embodiment, however, the former coupling order can be usedbecause the gain control is unnecessary.

As described above, the power supply device of FIG. 1 mainly has the twofeatures (square circuit and power supply topology), but is notnecessarily required to have both features. However, even in the case ofthe provision of only either one of the two features, a sufficientuseful advantage is obtained. The power supply device may preferablyhave both features to obtain a more beneficial advantage.

Second Embodiment

A second embodiment will explain a modification of the power supplydevice shown in FIG. 1.

<<Overall Circuit Configuration of Power Supply Device [2]>>

FIG. 11 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a second embodimentof the present invention. A description will now be made while payingattention to points of difference between the present power supplydevice and the power supply device shown in FIG. 1. The power supplydevice shown in FIG. 11 is different from the power supply device ofFIG. 1 in that the point of coupling of a ground power supply voltageGND1 of a PFC circuit and the point of acquisition of a detectionvoltage Vcs with it, and the position of a diode D1 differ, and that anoutput voltage Vout is fed back to the PFC circuit. That is, in thepower supply device of FIG. 11, the ground power supply voltage GND1 ofthe PFC circuit PIC3 is coupled to a node Nsw on the transistor Q1 sideof a resistor Rcs, and the PFC circuit PIC3 acquires a detection voltageVcs from a node on the inductor L1 side of the resistor Rcs. The diodeD1 is inserted between one end of the inductor L1 and a positivepolarity output node Vout (+) with the L1 side as its anode and the Vout(+) side as the cathode. With its insertion, a negative polarity outputnode Vout (−) is coupled to the node Nsw and the ground power supplyvoltage GND1.

Further, the power supply device shown in FIG. 11 includes resistorsRfb1 and Rfb2 coupled in series between the positive polarity outputnode Vout (+) and the negative polarity output node Vout (−) and outputsan output voltage detection signal Vout′ from a node for resistancedivision of the output voltage Vout by the resistors. The output voltagedetection signal Vout′ is inputted to an error amplifier circuit EA ofthe PFC circuit PIC3. The error amplifier circuit EA detects adifference between the output voltage detection signal Vout′ and a setvalue (comparison voltage Vr2) of the output voltage Vout determined inadvance and outputs the same to a multiplication circuit MUL. A signalVm outputted from the multiplication circuit MUL is inputted to acomparator circuit CMPp via a square circuit SQ in a manner similar tothe PFC circuit PIC1 of FIG. 1. The PFC circuit PIC3 has an inversebuffer circuit IBF which receives a detection voltage Vcs developed bythe resistor Rcs therein as an input and outputs the same at a-1-timegain. The output voltage of the inverse buffer circuit IBF is comparedwith the output signal Vs of the square circuit SQ by the comparatorcircuit CMPp and compared with a comparison voltage Vr1 by a comparatorcircuit CMPz.

Assuming now where the voltage feedback system is applied to the powersupply device of FIG. 1 (i.e., where such resistors Rfb1 and Rfb2 asshown in FIG. 11 are added), the PFC circuit detects a voltage obtainedby adding a voltage developed by the resistor Rcs and the diode D1 tothe output voltage detected by resistance division by the resistors Rfb1and Rfb2. In this case, there is a fear that an error occurs in thedetected value of the output voltage. On the other hand, when the powersupply device shown in FIG. 11 is used, the PFC circuit PIC3 detects anoutput voltage with Vout (−) as GND1, thereby making it possible todetect the output voltage with a high degree of accuracy.

Third Embodiment

A third embodiment will explain a case in which the square circuit SQdescribed in the first embodiment is applied to a flyback converter.

<<Overall Circuit Configuration of Power Supply Device [3]>>

FIG. 12 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a third embodimentof the present invention. The power supply device shown in FIG. 12includes a rectifying circuit DB1, a power factor correction (PFC)circuit (semiconductor device) PIC4, a transistor (switch element) Q1,resistors Rac1, Rac2, Rcs, Rfb1 and Rfb2, a transformer TR1, a diode D1,and an output capacitor Cout. The output capacitor Cout is coupledbetween a positive polarity output node Vout (+) and a negative polarityoutput node Vout (−) and supplies an output voltage Vout generated atthe positive polarity output node Vout (+) with the negative polarityoutput node Vout (−) as a reference to a load circuit (not shown). Thenegative polarity output node Vout (−) is coupled to a ground powersupply voltage GND1.

The rectifying circuit DB1 full-wave rectifies a commercial power source(AC) (ac voltage Vac and ac current Iac) and outputs an input voltageVin and an input current Iin to a rectified output node Ndb1 with theground power supply voltage GND1 as the reference. The resistors Rac1and Rac2 are coupled in series between the rectified output node Ndb1and the ground power supply voltage GND1 and outputs, from a commoncoupling node thereof, an input voltage detection signal Vin′ thatassumes a value generated by resistance voltage division from the inputvoltage Vin. The transformer TR1 has a primary winding (inductor) LM1, asecondary winding (inductor) LM2 and a supplementary winding (inductor)LMs. The transistor (n channel-type power MOS transistor herein) Q1 hasa drain coupled to the rectified output node Ndb1 via the primarywinding LM1, a source coupled to one end of the resistor Rcs, and a gatedriven by a duty control signal PWM generated from the PFC circuit PIC4.The other end of the resistor Rcs is coupled to the ground power supplyvoltage GND1. The diode D1 has an anode coupled to the ground powersupply voltage GND1 via the secondary winding LM2, and a cathode coupledto the positive polarity output node Vout (+). The resistors Rfb1 andRfb2 are coupled in series between the positive polarity output nodeVout (+) and the negative polarity output node Vout (−) (GND1) andoutputs, from a common coupling node thereof, an output voltagedetection signal Vout′ that assumes a value generated by resistancevoltage division from the output voltage Vout.

The PFC circuit (semiconductor device) PIC4 is operated by the groundpower supply voltage GND1 and a power supply voltage VCC and outputs aduty control signal PWM in response to the input voltage detectionsignal Vin′, a detection voltage Vcs by the resistor Rcs and the outputvoltage detection signal Vout′. The PFC circuit PIC4 includes amultiplication circuit MUL, a square circuit SQ, comparator circuitsCMPp and CMPz, a set/reset latch circuit SRLT, a driver circuit DRV, anerror amplifier circuit EA and a protection zener diode ZD1.

When a reset signal RT is outputted from the comparator circuit CMPp,the set/reset latch circuit SRLT drives the duty control signal PWM toan ‘L’ level (off level) through the driver circuit DRV. When a setsignal ST is outputted from the comparator circuit CMPz, the set/resetlatch circuit SRLT drives the duty control signal PWM to an ‘H’ level(on level) through the driver circuit DRV. The comparator circuit CMPzoutputs the set signal ST when a detection voltage Vz obtained by thesupplementary winding LMs of the transformer TR1 becomes lower than apredetermined comparison voltage Vr1. Incidentally, when a high voltageoccurs in the detection voltage Vz, the comparator circuit CMPz isprotected by a clamp operation of the zener diode ZD1. The erroramplifier circuit EA amplifies a difference between the output voltagedetection signal Vout′ and a predetermined comparison voltage Vr2. Themultiplication circuit MUL multiplies the output voltage of the erroramplifier circuit EA and the input voltage detection signal Vin′ by eachother and thereby generates an output signal Vm indicative of the resultof its multiplication. The square circuit SQ squares the output signalVm and generates an output signal Vs indicative of the result of itssquare. The comparator circuit CMPp outputs a reset signal RT when thedetection voltage Vcs based on the resistor Rcs exceeds the outputsignal Vs.

<<Overall Circuit Operation of Power Supply Device [3]>>

FIG. 13 is a waveform diagram showing a schematic example of operationof FIG. 12. Firstly, when a duty control signal PWM is at an ‘H’ level(on level) as shown in FIG. 13, the transistor Q1 is driven on, so thatan increasing input current Iin flows through the primary winding LM1 ofthe transformer TR1 and power is accumulated in the transformer TR1.With an increase in the input current Iin, the corresponding detectionvoltage Vcs also increases in growing form. On the other hand, when thedetection voltage Vs reaches a voltage value of an output signal Vs ofthe square circuit SQ, a reset signal RT is generated from thecomparator circuit CMPp and hence the duty control signal PWM istransitioned to an ‘L’ level (off level), so that the transistor Q1 isdriven off. Since the voltage polarities of the primary and secondarywindings LM1 and LM2 of the transformer TR1 are respectively inverted indoing so, the power accumulated in the transformer TR1 is discharged viathe secondary winding LM2. That is, the diode D1 is biased in theforward direction so that an output current Iout is supplied to thecorresponding load circuit and thereby the charging of the outputcapacitor Cout is performed. During a period in which the transistor Q1is off, the current (output current Iout) flowing through the secondarywinding LM2 is reduced in decreasing form. Then, when the poweraccumulated in the transformer TR1 reaches zero (Iout reaches zero), avoltage Vz detected at the supplementary winding LMs suddenly drops tozero. When the voltage Vz falls below the comparison voltage Vr1 of thecomparator circuit CMPz, a set signal ST is generated so that thetransistor Q1 is driven on again. That is, the power supply device ofFIG. 12 operates in a current critical mode.

In FIG. 12, the error amplifier circuit EA detects a difference betweenan output voltage detection signal Vout′ and a set value (Vr2) of apredetermined Vout′. The multiplication circuit MUL reflects the resultof detection by the error amplifier circuit EA on the input voltagedetection signal Vin′ to generate an output signal Vm. Since the inputvoltage detection signal Vin′ takes a waveform proportional to an inputvoltage Vin that becomes a sinusoidal wave (strictly the absolute valuewaveform of sinusoidal wave), the output signal Vm assumes the sine wave(strictly the absolute value waveform of sinusoidal wave), and thevoltage amplitude thereof becomes a waveform that varies according tothe result of detection by the error amplifier circuit EA. Since theoutput signal Vs of the square circuit SQ is of a value obtained bysquaring the output signal Vm, the output signal assumes such a waveshape that the sinusoidal wave is pointed as shown in FIG. 13.

Here, in a manner similar to the power supply device of FIG. 1, thepower supply device shown in FIG. 12 takes a circuit system in which theinput current Iin flows during the on period of the trnasistor Q1 andthe output current Iout flows during its off period. If the squarecircuit SQ is not provided in this case, then the waveform of the accurrent Iac (input current Iin) can take the trapezoidal shape asdescribed in FIG. 34. Thus, the period of the ‘H’ level (on oftransistor Q1) of the duty control signal PWM is determined based on theoutput signal Vs having such a wave shape that the sinusoidal wave ispointed, thereby making it possible to bring the ac current Iacdeveloped in the commercial power line (AC) close to the sinusoidalwave.

<<Detailed Circuit Operation of Power Supply Device [3]>>

A detailed example of operation of the principal part in the powersupply device of FIG. 12 will next be explained with reference to FIG.14. FIG. 14 is a waveform diagram showing a part of FIG. 13 in developedform and shows a duty control signal (gate voltage of transistor Q1)PWM, and an input current Iin flowing through the primary side of thetransformer TR1 and an output current Iout flowing through the secondaryside thereof, all of which are extracted in FIG. 13. First assume thatthe inductance value of the primary winding LM1 is LM1, the inductancevalue of the secondary winding LM2 is LM2, the number of turns in theprimary winding LM1 is n1, and the number of turns in the secondarywinding LM2 is n2, the relationship of the following equation (1C) isestablished in terms of the characteristics of the transformer:

n1×Iin=n2×Iout

LM2=LM1×(n2/n1)²   (1C)

Referring to FIG. 14, the peak value Ipk1 of the input current Iin isexpressed in the following equation (2C) using an on period Ton of thetransistor Q1, and the peak value Ipk2 of the output current Iout isexpressed in the following equation (3C) using an off period Toff of thetransistor Q1 and an output voltage Vout.

Ipk1=(Vin/LM1)·Ton   (2C)

Ipk2=(Vout/LM2)·Toff   (3C)

Substituting the equations (1C) and (2C) into the equation (3C) yieldsthe following equation (4C), and one cycle Tsw of the duty controlsignal PWM is expressed in the following equation (5C).

$\begin{matrix}{{Toff} = {\left( {{Vin}/\left( {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}} \right)} \right) \times {Ton}}} & \left( {4C} \right) \\{{Tsw} = {{{Ton} + {Toff}} = {\frac{{Vin} + {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}}}{\left( {n\; {1/n}\; 2} \right) \cdot {Vout}} \times {Ton}}}} & \left( {5C} \right)\end{matrix}$

Using these equations, the average value Iin_ave of the input currentIin in one cycle of the duty control signal PWM is defined by thefollowing equation (6C):

$\begin{matrix}{{I\; {in\_ ave}} = {\frac{\left( {1/2} \right) \times {Ipk}\; 1 \times {Ton}}{Tsw} = {{1/\left( {{2 \cdot L}\; M\; 1} \right)} \times \frac{{Vin} \times {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}}}{{Vin} + {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}}} \times {Ton}}}} & \left( {6C} \right)\end{matrix}$

On the other hand, when a resistance division ratio of the resistorsRac1 and Rac2 to the input voltage Vin is assumed to be K2, the inputvoltage detection signal Vin′ becomes Vin′=K2×Vin. When the outputsignal of the error amplifier circuit EA is assumed to be K1, the outputsignal Vm of the multiplication circuit MUL becomes Vm=K1·K2·Vin.Accordingly, the output signal Vs of the square circuit SQ is expressedin the following equation (7C) as K=(K1·K2)²:

Vs=K·Vin²   (7C)

Since the comparator circuit CMPp controls the transistor Q1 to off whenthe detection voltage Vcs based on the resistor Rcs reaches the outputsignal Vs, Rcs×Ipk1 is set to Rcs×Ipk1=equation (7C) using Ipk1 in theequation (2C). As a result, Ton is expressed in the following equation(8C):

Ton=((K·M1)/Rcs)−Vin   (8C)

Substituting the equation (8C) into the equation (6C) yields thefollowing equation (9C) as the average value Iin_ave:

$\begin{matrix}{{I\; {in\_ ave}} = {{\frac{K}{2 \times R\; {cs}} \cdot \frac{{Vin} \times {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}}}{{Vin} + {\left( {n\; {1/n}\; 2} \right) \cdot {Vout}}}} \times {Vin}}} & \left( {9C} \right)\end{matrix}$

Thus, the power supply device (PFC circuit PIC4) shown in FIG. 12acquires the input voltage detection signal Vin′ in the form of afunction of Vin under the concept of the numerical expression andsquares the same to generate the term of Vin in the Ton as shown in theequation (8C), and cancels out the value (Vin+(n1/n2)·Vout) of thedenominator in the equation (6C) by the Ton. Thus, the average valueIin_ave of the equation (6C) is brought close to the function of theinput voltage Vin corresponding to the sinusoidal wave.

On the other hand, since the average value Iin_ave is set as the perfectfunction of Vin, there is also considered a further configurationexample provided with an adder, a constant generator, etc. In doing so,there is a fear that since additional external terminals are requiredand the circuit area of the PFC circuit increases in a manner similar tothe case of FIG. 6, a size reduction in the power supply device is notsufficiently achieved. Thus, when such a power supply device as shown inFIG. 12 is used, a substantially sinusoidal-shaped input currentwaveform which is not a perfect sinusoidal wave but has sufficientquantity in practical use (is capable of reducing higher harmoniccomponents as required and sufficiently), can be generated owing to itssmall-sized and low-cost configuration.

FIGS. 15(a) and 15(b) are respectively waveform diagrams showing resultsof simulation of input current waveforms where the power supply deviceshown in FIG. 12 is used. One example illustrative of an ideal inputcurrent waveform (ideal system) indicative of a sinusoidal wave and atheoretical input current waveform (square circuit control) calculatedby the equation (9C) where the power supply device of FIG. 12 is used,is shown in FIG. 15(a). One example indicative of an ideal input currentwaveform (with no square circuit) calculated by the equation (2A) wherethe above power supply device of FIG. 31 is used, is shown in FIG. 15(b)as a target for comparison thereof. As is understood from the comparisonbetween FIGS. 15(a) and 15(b), the use of the power supply device ofFIG. 12 enables the input current waveform to be brought close to thesine wave, and higher harmonic components can be reduced sufficientlyfrom a practical standpoint.

<<Various Modifications of Power Supply Device [3]>>

FIGS. 16 through 18 are respectively schematic diagrams showing oneexample illustrative of different circuit configurations in which thepower supply device of FIG. 12 is modified. A description will now bemade while paying attention to the differences between the power supplydevice of FIG. 12 and these different power supply devices. The powersupply device of FIG. 16 is different from the power supply device ofFIG. 12 in terms of an output information feedback method. That is, thepower supply device of FIG. 12 is set as the configuration example inwhich information (i.e., output voltage detection signal Vout′) aboutthe output voltage Vout is fed back to the PFC circuit PIC4, whereas thepower supply device of FIG. 16 is set as a configuration example inwhich information about an output current Iout is fed back to a PFCcircuit PIC5 as an output current detection signal Iout′. For example,the power supply device shown in FIG. 12 is of a configuration exampleapplied when voltage-driven load circuits are combined. The power supplydevice shown in FIG. 16 is of a configuration example applied whencurrent-driven load circuits such as light emitting diodes or the likeare combined.

In FIG. 16, the output current Iout is converted to its correspondingvoltage by a resistor Rfb3 inserted onto a current path (between Vout(−) and GND1) of a negative polarity output node Vout (−). An erroramplifier circuit EA provided outside the PFC circuit PIC5 detects adifference between a voltage indicative of the magnitude of Ioutdeveloped at one end (Vout (−)) of the resistor Rfb3 and a set value(comparison voltage Vr2) of Iout determined in advance. Then, the baseof a pnp type bipolar transistor QP20 provided outside the PFC circuitPIC5 is controlled by the result of detection so that an output currentdetection signal Iout′ is generated from its emitter.

On the other hand, the PFC circuit PIC5 is provided with a wiring nodeNiof for coupling one of two inputs in a multiplication circuit MUL tothe emitter of the pnp type bipolar transistor QP20, and a constantcurrent source IBS which supplies current for determining an operatingpoint of the pnp type bipolar transistor QP20, via the wiring node Niof,instead of the error amplifier circuit EA provided in the PFC circuitPIC4 of FIG. 12. The constant current source IBS and the pnp typebipolar transistor QP20 serve as a variable current source on which themagnitude of the output current Iout is reflected. This variable currentsource may also be used as the constant current source IS2 of themultiplication circuit MUL shown in FIG. 9.

The power supply device shown in FIG. 17 is of a configuration examplein which the non-insulation type power supply device shown in FIG. 12 ischanged to an insulation type. Likewise, the power supply device of FIG.18 is of a configuration example in which the non-insulation type powersupply device shown in FIG. 16 is changed to the insulation type. Forexample, there is a case in which a power supply device targeted for aload circuit large in output power needs to have a non-insulation typeconfiguration from the viewpoint of safety or the like. In such a case,such a power supply device as shown in FIG. 17 or 18 may be used.

In FIG. 17, the primary winding LM1 side of a transformer TR1 isoperated on the basis of a ground power supply voltage GND1, whereas thesecondary winding LM2 side thereof is operated on the basis of a groundpower supply voltage GND2. That is, a negative polarity output node Vout(−) is coupled to the GND2. An output voltage Vout generated at apositive polarity output node Vout (+) on the basis of the negativepolarity output node Vout (−) is divided by resistors Rfb1 and Rfb2. Anode for this resistance division is coupled to a control node of ashunt regulator SR1. A resistor Rfb4 and an input path of a photocouplerPC1 are coupled in series between the cathode of the shunt regulator SR1and the positive polarity output node Vout (+) in order from thepositive polarity output node Vout (+) side. The shunt regulator SR1varies the voltage of the cathode thereof in such a manner that thevoltage of its control node becomes a reference voltage Vref set insidethe shunt regulator SR1 (i.e., Vout=Vref×(Rfb1+Rfb2)/Rfb2). A currentflowing through the input path of the photocoupler PC1 variescorrespondingly. Thus, a current on which such current is reflected, isextracted or taken out from an output path of the photocoupler PC1 andconverted via a resistor Rfb5 to a voltage as an output voltagedetection signal Vout′. The PFC circuit PIC4 is similar in configurationto FIG. 12 and operates in response to the output voltage detectionsignal Vout′.

Even in FIG. 18 as with the case of FIG. 17, the primary winding LM1side of a transformer TR1 is operated on the basis of a ground powersupply voltage GND1, whereas the secondary winding LM2 side of thetransformer TR1 is operated on the basis of a ground power supplyvoltage GND2. An output current Iout flowing through the secondarywinding LM2 side is converted into a voltage by a resistor Rfb3 insertedonto a current path of a negative polarity output node Vout (−) (betweenVout (−) and GND2). An error amplifier circuit EA detects a differencebetween a voltage indicative of the magnitude of Iout developed at oneend (Vout (−)) of the resistor Rfb3 and a set value (comparison voltageVr2) of Iout determined in advance, and controls current flowing throughan input path of a photocoupler PC1 according to the result ofdetection. Then, a current on which the current flowing through theinput path of the photocoupler PC1 is reflected, is extracted or takenout from an output path of the photocoupler PC1. The PFC circuit PIC5 issimilar in configuration to FIG. 16 and operates in response to theoutput current Iout′.

Thus, higher harmonic components likely to occur in the input currentcan be reduced by using the power supply device according to the thirdembodiment, typically, by the small-sized or low-cost configuration.This advantageous effect is obtained by using the square circuit SQ inthe PFC circuit, instead of the use of a complex circuit in which anadder and the like are combined. Since the corresponding power supplydevice needs to have the transformer as compared with the power supplydevice according to the first embodiment, the power supply deviceaccording to the first embodiment is desired in terms of a sizereduction or a reduction in cost. However, when the insulation typepower supply device is required, for example, the use of the powersupply device according to the third embodiment becomes useful.

Fourth Embodiment

The first embodiment has shown the configuration example in which thesquare circuit SQ is applied to the high-side inverse converter in whichthe transistor Q1 is disposed on the high-potential output side of therectifying circuit DB1, and the inductor L1 is disposed on thelow-potential output side. A fourth embodiment will explain aconfiguration example in which a square circuit SQ is applied to alow-side inverse converter in which an inductor is disposed on thehigh-potential output side of a rectifying circuit, and a transistor isdisposed on the low-potential output side.

<<Overall Circuit Configuration of Power Supply Device [4]>>

FIG. 19 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a fourth embodimentof the present invention. The power supply device shown in FIG. 19 has arectifying circuit DB1, a power factor correction (PFC) circuit(semiconductor device) PIC4, a transformer TR2, a transistor Q1, a diodeD1, an output capacitor Cout, resistors Rac1, Rac2, Rcs, Rfb11 throughRfb14, and an operational amplifier OP1. The output capacitor Cout iscoupled between a positive polarity output node Vout (+) and a negativepolarity output node Vout (−) and supplies an output voltage Voutgenerated at the positive polarity output node Vout (+) on the basis ofthe negative polarity output node Vout (−) to a load circuit (notshown).

The rectifying circuit DB1 full-wave rectifies a commercial power source(AC) (ac voltage Vac and ac current Iac) and outputs an input voltageVin and an input current Iin to a rectified output node Ndb1 on thebasis of the ground power supply voltage GND1. The resistors Rac1 andRac2 are coupled in series between the rectified output node Ndb1 andthe ground power supply voltage GND1 and outputs, from a common couplingnode thereof, an input voltage detection signal Vin′ that assumes avalue generated by resistance voltage division from the input voltageVin. The transformer TR2 has an inductor LM1 and a supplementary winding(inductor) LMs. The transistor (n channel-type power MOS transistorherein) Q1 has a drain coupled to the rectified output node Ndb1 via theinductor LM1, a source coupled to one end of the resistor Rcs, and agate driven by a duty control signal PWM generated from the PFC circuitPIC4. The other end of the resistor Rcs is coupled to the ground powersupply voltage GND1. The diode D1 has an anode coupled to the drain ofthe transistor Q1 and a cathode coupled to the positive polarity outputnode Vout (+).

The resistor Rfb11 is coupled between the negative polarity output nodeVout (−) and a negative polarity input node of the operational amplifierOP1, and the resistor Rfb12 is coupled between the positive polarityoutput node Vout (+) and a positive polarity input node of theoperational amplifier OP1. The resistor Rfb13 is coupled between thenegative polarity input node of the operational amplifier OP1 and anoutput node of the operational amplifier OP1, and the resistor Rfb14 iscoupled between the positive polarity input node of the operationalamplifier OP1 and the ground power supply voltage GND1. Thus, theoperational amplifier OP1 serves as a differential amplifier circuit andgenerates an output voltage detection signal Vout′ from its output node.When the resistance values of the resistors Rfb11 and Rfb12 are bothassumed to be R11, and the resistance values of the resistors Rfb13 andRfb14 are both assumed to be R13, Vout′=(R13/R11)×(Vout (+)−Vout (−)).Incidentally, the PFC circuit PIC4 is similar in configuration to thatin FIG. 12 and outputs a duty control signal PWM in response to adetection voltage Vcs outputted from a coupling node of the transistorQ1 and the resistor Rcs, and a detection voltage Vz generated by thesupplementary winding LMs.

<<Overall Circuit Operation of Power Supply Device [4]>>

When the duty control signal PWM is at an ‘H’ level (on level) in such aconfiguration, the transistor Q1 is driven on, so that a growing inputcurrent Iin flows through the inductor LM1 of the transformer TR2 andpower is accumulated in the inductor LM1. With an increase in the inputcurrent Iin, the corresponding detection voltage Vcs based on theresistor Rcs also increases in growing form. On the other hand, when thedetection voltage Vs reaches a voltage value of an output signal Vs ofthe square circuit SQ, a reset signal RT is generated from thecomparator circuit CMPp and hence the duty control signal PWM istransitioned to an ‘L’ level (off level), so that the transistor Q1 isdriven off. In doing so, with the accumulated power of the LM1 as anelectromotive force, an output current Iout flows through a path of thediode D1, Vout (+) and Vout (−), and the charging of the outputcapacitor Cout is performed thereby. During a period in which thetransistor Q1 is off, the output current Iout is reduced in decreasingform. Then, when the output current Iout reaches zero, a voltage Vzdetected at the supplementary winding LMs suddenly drops to zero. Whenthe voltage Vz falls below the comparison voltage Vr1 of the comparatorcircuit CMPz, a set signal ST is generated so that the transistor Q1 isdriven on again. That is, the power supply device of FIG. 19 operates ina current critical mode.

In FIG. 19, the output voltage Vout between the positive polarity outputnode Vout (+) and the negative polarity output node Vout (−) is fed backto the PFC circuit PIC4 as an output voltage detection signal Vout′ bythe differential amplifier circuit comprised of OP1 or the like. At thistime, the differential amplifier circuit is provided to allow the PFCcircuit PIC4 to detect the output voltage Vout between the negativepolarity output node Vout (−) and the positive polarity output node Vout(+) on the basis of the ground power supply voltage GND1. By setting theratio (R13/R11) between the resistance value R11 and the resistancevalue R13 to a few tenths, for example, an output voltage detectionsignal Vout′ varied in a voltage range of a few V on the basis of theground power supply voltage GND1 can be generated according to the level(e.g., a few tens of V or so) of the output voltage Vout. In the PFCcircuit PIC4, its error amplifier circuit EA amplifies a differencebetween the output voltage detection signal Vout′ and a comparisonvoltage Vr2 (e.g., a few V or so). A multiplication circuit MUL reflectsthe result of detection by the error amplifier circuit EA on the inputvoltage detection signal Vin′ to generate an output signal Vm. Thesquare circuit SQ generates an output signal Vs in which the outputsignal Vm is squared.

Here, in a manner similar to the power supply devices according to thefirst through third embodiments, the power supply device shown in FIG.19 takes a circuit system in which the input current Iin flows duringthe on period of the trnasistor Q1 and the output current Iout flowsduring its off period. If the square circuit SQ is not provided in thiscase, then the waveform of the input current Iin can take thetrapezoidal shape. Thus, in a manner similar to the case of FIG. 1 orthe like, an output signal Vs having such a wave shape that a sinusoidalwave is pointed is generated by the square circuit SQ, and the period ofthe ‘H’ level (on of transistor Q1) of the duty control signal PWM isdetermined based on the output signal Vs, thereby making it possible tobring the ac current Iac developed in the commercial power line (AC)close to the sinusoidal wave.

<<Modification of Power Supply Device [4]>>

FIG. 20 is a schematic diagram showing one example of a modified circuitconfiguration of the power supply device of FIG. 19. A description willnow be made while paying attention to the differences between thepresent modification and the power supply device of FIG. 19. The powersupply device of FIG. 20 is different from the power supply device ofFIG. 19 in terms of an output information feedback method. That is, thepower supply device of FIG. 19 is illustrated as the configurationexample in which the information (i.e., output voltage detection signalVout′) about the output voltage Vout is fed back to the PFC circuitPIC4, whereas the power supply device of FIG. 20 takes a configurationexample in which information about an output current Iout is fed back toa PFC circuit PIC5 as an output current detection current Iout′.

In FIG. 20, the output current Iout is converted to its correspondingvoltage by a resistor Rfb3 inserted onto a current path (between one endof an output capacitor Cout and a negative polarity output node Vout(−)) of the negative polarity output node Vout (−). An error amplifiercircuit EA provided outside the PFC circuit PIC5 detects a differencebetween a voltage indicative of the magnitude of Iout developed at oneend (Vout (−)) of the resistor Rfb3 and a set value (comparison voltageVr2) of Iout determined in advance. Then, the base of a pnp type bipolartransistor QP40 provided outside the PFC circuit PIC5 is controlled bythe result of detection. The pnp type bipolar transistor QP40 has anemitter coupled to a positive polarity output node Vout (+) via aresistor Rfb21. A collector current flowing therethrough is inputted tothe collector (base) of an npn type bipolar transistor QN40. The npntype bipolar transistor QN40 configures a current mirror circuit betweenitself and an npn type bipolar transistor QN41. The current inputted tothe npn type bipolar transistor QN40 is transferred to the npn typebipolar transistor QN41. The current transferred to the npn type bipolartransistor QN41 is fed back to the PFC circuit PIC5 as an output currentdetection signal Iout′.

When the error amplifier circuit EA is operated with the Vout (−) as aground power supply voltage, for example, the output voltage of theerror amplifier circuit EA does not correspond to a suitable voltagerange as viewed from the ground power supply voltage GND1 of the PFCcircuit PIC5. For this reason, it is difficult to feed back the currentto the PFC circuit PIC5 via one bipolar transistor as in the case ofFIG. 16. Thus, a current signal corresponding to the output current Ioutis generated by the pnp type bipolar transistor QP40 and turned back bythe current mirror circuit, followed by being fed back to the PFCcircuit PIC5. The PFC circuit PIC5 is similar in configuration to thatof FIG. 16 and outputs a duty control signal PWM in response to theinput voltage detection signal Vin′, detection voltage Vz developed by asupplementary winding LMs, detection voltage Vcs produced by a resistorRcs, and output current detection signal Iout′.

Thus, higher harmonic components likely to occur in the input currentcan be reduced by using the power supply device according to the fourthembodiment, typically, by the small-sized or low-cost configuration.This advantageous effect is obtained by using the square circuit SQ inthe PFC circuit, instead of the use of a complex circuit in which anadder and the like are combined. However, since the corresponding powersupply device needs to have the transformer as compared with the powersupply device according to the first embodiment, the power supply deviceaccording to the fourth embodiment is desired in terms of a sizereduction or a reduction in cost.

Fifth Embodiment

While the above fourth embodiment has shown the configuration example inwhich the square circuit SQ is applied to the low-side inverseconverter, and the control in the current critical mode is performedusing the supplementary winding LMs, a fifth embodiment will explain aconfiguration example in which control in a current critical mode isperformed using a resistor instead of the supplementary winding.

<<Overall Circuit Configuration of Power Supply Device [5]>>

FIG. 21 is a schematic diagram depicting one example of a circuitconfiguration of a power supply device according to the fifth embodimentof the present invention. A description will now be made while payingattention to the differences between the present power supply device andthe power supply device of FIG. 19. The power supply device shown inFIG. 21 has a configuration in which the transformer TR2 shown in FIG.19 is replaced with an inductor L1 and a resistor Rcs2 is providedbetween the inductor L1 and a positive polarity output node Vout (+), ascompared with the power supply device of FIG. 19. When an output currentIout flowing through the inductor L1 reaches zero where a transistor Q1is off, a detection voltage Vz developed across the resistor Rcs2 alsoreaches zero. Therefore, it is detected by a comparator circuit CMPzprovided within a PFC circuit PIC4′ to thereby enable the control in thecurrent critical mode. Incidentally, the PFC circuit PIC4′ is differentfrom the PFC circuit PIC4 of FIG. 19 in terms of an input part of thecomparator circuit CMPz. That is, the detection voltage Vz developed inthe supplementary winding on the basis of the ground power supplyvoltage GND1, and the comparison voltage Vr1 set on the basis of theground power supply voltage GND1 are inputted to the comparator circuitCMPz of the PFC circuit PCI4. In contrast, a detection voltage Vzdeveloped in the resistor Rcs2 on the basis of the positive polarityoutput node Vout (+), and a comparison voltage Vr1 set on the basis ofthe positive polarity output node Vout (+) are inputted to thecomparator circuit CMPz of the PFC circuit PIC4′.

<<Modification of Power Supply Device [5]>>

FIG. 22 is a schematic diagram showing one example of a modified circuitconfiguration of the power supply device of FIG. 21. The power supplydevice shown in FIG. 22 is illustrated as a configuration example inwhich output current information is fed back to a PFC circuit PIC5′while the power supply device shown in FIG. 21 is illustrated as theconfiguration example in which the output voltage information is fedback to the PFC circuit PIC4′. That is, the power supply device shown inFIG. 22 is provided with a resistor Rcs2 for operation in a currentcritical mode as a comparative target for the power supply device ofFIG. 20 in a manner similar to the case of the power supply device ofFIG. 21. Further, the power supply device of FIG. 22 is equivalent toone in which the input part of the comparator circuit CMPz in the PFCcircuit PIC5 of FIG. 20 is changed as with the power supply device ofFIG. 21.

Since the use of the power supply device according to the fifthembodiment makes it unnecessary to provide the supplementary winding(i.e., transformer) as compared with the power supply device accordingto the fourth embodiment, a further size reduction in the power supplydevice, and the like can be achieved. Since, however, a voltage level(may be 100 V or higher assuming that the ground power supply voltageGND1 is taken as the reference) developed at the positive polarityoutput node Vout (+) is inputted to each of the PFC circuits PIC4′ andPIC5′ operated with the GND1 as the reference, it is necessary to ensurethe breakdown voltages of the PFC circuits PIC4′ and PIC5′. Accordingly,the power supply device according to the first embodiment is preferablyused from this point of view.

Sixth Embodiment

A sixth embodiment will explain a power supply device which performs anoperation in a current continuous mode other than the current criticalmode using the power supply topology of the high-side inverse converterin a manner similar to the first and second embodiments.

<<Overall Circuit Configuration of Power Supply Device [6]>>

FIG. 23 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a sixth embodimentof the present invention. A description will now be made while payingattention to the differences between the present power supply device andthe power supply device of FIG. 11 described in the second embodiment.The power supply device shown in FIG. 23 is different from the powersupply device of FIG. 11 in terms of the position of a resistor Rcs andan internal circuit configuration of a PFC circuit PIC6. One end of theresistor Rcs is coupled to a ground power supply voltage GND1 of the PFCcircuit PIC6 in a manner similar to FIG. 11, but the other end of theresistor Rcs is coupled to a transistor Q1 unlike FIG. 11. A detectionvoltage Vcs is outputted from a node on the transistor Q1 side, of theresistor Rcs. With it, one end of an inductor L1 is coupled to theground power supply voltage GND1 and negative polarity output node Vout(−).

The PFC circuit (semiconductor device) PIC6 is operated by the groundpower supply voltage GND1 and power supply voltage VCC, and outputs aduty control signal PWM in response to an input voltage detection signalVin′, a detection voltage Vcs obtained by the resistor Rcs and an outputvoltage detection signal Vout′. The PFC circuit PIC6 includes amultiplication circuit MUL, two error amplifier circuits EA1 and EA2, acomparator circuit CMPp, a set/reset latch circuit SRLT, a drivercircuit DRV, an amplifier circuit AMPi, an oscillator OSC, and atriangular wave generator TWGEN.

The set/reset latch circuit SRLT drives the duty control signal PWM toan ‘L’ level (off level) via the driver circuit DRV when a reset signalRT is outputted from the comparator circuit CMPp, and drives the dutycontrol signal PWM to an ‘H’ level (on level) via the driver circuit DRVwhen a set signal ST is outputted from the oscillator OSC. The amplifiercircuit AMPi amplifies the detection voltage Vcs and outputs the same toone of two inputs of the error amplifier circuit EA2. The erroramplifier circuit EA1 amplifies the output voltage detection signalVout′ detected by the resistors Rfb1 and Rfb2 on the basis of acomparison voltage Vr2. The multiplication circuit MUL multiplies theinput voltage detection signal Vin′ and the output voltage of the erroramplifier circuit EA1 by each other and outputs the result ofmultiplication to the other of the two inputs of the error amplifiercircuit EA2. The error amplifier circuit EA2 amplifies a differencebetween the output voltage of the amplifier circuit AMPi and the outputvoltage of the multiplication circuit MUL and outputs the same to one oftwo inputs of the comparator circuit CMPp. The triangular wave generatorTWGEN generates a triangular wave signal using a signal outputted fromthe oscillator OSC and outputs the same to the other of the two inputsof the comparator circuit CMPp.

<<Overall Circuit Operation of Power Supply Device [6]>>

FIG. 24 is a waveform diagram showing a schematic example of operationof FIG. 23. Firstly, when the oscillator OSC outputs a set signal ST, aduty control signal PWM is brought to an ‘H’ level (on level), so thatthe transistor Q1 is driven on. Thus, a growing input current Iin flowsthrough the inductor L1 via the transistor Q1 and the resistor Rcs, sothat power is accumulated in the inductor L1. With an increase in theinput current Iin, the corresponding detection voltage Vcs developed atone end of the resistor Rcs also increases in growing form. Thereafter,when a reset signal ST is outputted from the comparator circuit CMPp,the duty control signal PWM is brought to an ‘L’ level (on level), sothat the transistor Q1 is driven off. In doing so, with the poweraccumulated in the L1 as an electromotive force, an output current Ioutflows through a path of a positive polarity output node Vout (+), adiode D1, and the negative polarity output node Vout (−). Thus, a loadcircuit (not shown) is driven by the output current Iout and an outputcapacitor Cout is charged thereby.

When the transistor Q1 is on, the error amplifier circuit EA1 outputs anerror component of the output voltage Vout relative to the set voltage(Vr2), and the multiplication circuit MUL generates an output voltage inwhich the error component is reflected on the input voltage detectionsignal Vin′. The error amplifier circuit EA2 amplifies a differencebetween the output voltage of the multiplication circuit MUL and theoutput voltage of the amplifier circuit AMPi. On the basis of the outputvoltage of the multiplication circuit MUL, the level of the outputvoltage of the error amplifier circuit EA2 increases when the outputvoltage of the amplifier circuit AMPi is low, so that a period (i.e., onduty of transistor Q1) from the output of the set signal ST to theoutput of the reset signal RT from the comparator circuit CMPpincreases. When the output voltage of the amplifier circuit AMPi islarge in reverse, the level of the output voltage of the error amplifiercircuit EA2 is lowered, so that the on duty of the transistor Q1 isreduced. Then, when the period corresponding to one cycle of theoscillator OSC has elapsed, the set signal ST is outputted again so thatthe transistor Q1 is driven on. With such a control operation, the inputcurrent Iin (ac current Iac that flows through the AC power line) thatflows through the transistor Q1 follows the input voltage Vin andassumes a sinusoidal shape.

<<Details of Oscillator and Triangular Wave Generator>>

FIGS. 25(a) and 25(b) show the details of the oscillator OSC andtriangular wave generator TWGEN in FIG. 23, in which FIG. 25(a) is acircuit diagram showing its configuration example, and FIG. 25(b) is awaveform diagram showing an example of operation of FIG. 25(a). Ahysteresis comparator circuit CMPh, an NMOS transistor MN1on/off-controlled according to the output of the hysteresis comparatorcircuit CMPh, a capacitor C1 coupled to a (+) input node of thehysteresis comparator circuit CMPh, a constant current source IScg whichcharges the capacitor C1, and a constant current source ISdcg whichperforms charging from the capacitor C1 via the NMOS transistor MN1.Here, the drive capacity of the constant current source ISdcg is setsufficiently higher than that of the constant current source IScg. Oneof a high-potential side threshold value Vh and a low-potential sidethreshold value V1 is set to a (−) input node of the hysteresiscomparator circuit CMPh. The high-potential side threshold value Vh isset when the hysteresis comparator circuit CMPh outputs an ‘L’ level,and the low-potential side threshold value V1 is set when the hysteresiscomparator circuit CMPh outputs an ‘H’ level.

When such a configuration example is used, the capacitor C1 is chargedfrom the constant current source IScg when the output of the hysteresiscomparator circuit CMPh is at the ‘L’ level, and the voltage level ofthe (+) input node of the hysteresis comparator circuit CMPh rises. Whenthe voltage level reaches Vh, the output of the hysteresis comparatorcircuit CMPh is transitioned to the ‘H’ level. With its transition, theelectric charge of the capacitor C1 is discharged via the constantcurrent source ISdcg, and the voltage level of the (+) input level ofthe hysteresis comparator circuit CMPh is hence lowered. When thevoltage level reaches V1, the output of the hysteresis comparatorcircuit CMPh is transitioned to the ‘L’ level again, and a similaroperation is repeated. As a result, an oscillation signal having apredetermined cycle is generated from the output of the hysteresiscomparator circuit CMPh, and a triangular wave signal is generated fromthe (+) input node of the comparator circuit CMPh. This oscillationsignal becomes an output signal of the oscillator OSC of FIG. 23, andthe triangular wave signal becomes an output signal of the triangularwave generator TWGEN of FIG. 23.

Seventh Embodiment

A seventh embodiment will explain a power supply device which performsan operation in a current discontinuous mode other than the currentcritical mode, using the power supply topology of the high-side inverseconverter as with the first and second embodiments.

<<Overall Circuit Configuration of Power Supply Device [7]>>

FIG. 26 is a schematic diagram illustrating one example of a circuitconfiguration of a power supply device according to a seventh embodimentof the present invention. A description will now be made while payingattention to the differences between the present power supply device andthe power supply device of FIG. 11 described in the second embodiment.The power supply device shown in FIG. 26 is different from the powersupply device shown in FIG. 11 in that the resistor Rcs for currentdetection and the resistors Rac1 and Rac2 for input voltage detectionare omitted, and an internal circuit configuration of a PFC circuit PIC7differs. The PFC circuit (semiconductor device) PIC7 operates inaccordance with a ground power supply voltage GND1 and a power supplyvoltage VCC and outputs a duty control signal PWM in response to anoutput voltage detection signal Vout′. The PFC circuit PIC7 includes anerror amplifier circuit EA1, a comparator circuit CMPp, a set/resetlatch circuit SRLT, an NOR arithmetic circuit NOR, a driver circuit DRV,an oscillator OSC, and a triangular wave generator TWGEN.

The set/reset latch circuit SRLT outputs an ‘H’ level to one of twoinputs of the NOR arithmetic circuit NOR when a set signal ST isoutputted from comparator circuit CMPp, and outputs an ‘L’ level to oneof the two inputs of the NOR arithmetic circuit NOR when a reset signalRT is inputted from the oscillator OSC. The output of the oscillator OSCis inputted to the other of the two inputs of the NOR arithmetic circuitNOR, and the output of the NOR arithmetic circuit NOR is outputted asthe duty control signal PWM via the driver circuit DRV. The erroramplifier circuit EA1 amplifies a difference between the output voltagedetection signal Vout′ and a comparison voltage Vr2 and outputs the sameto one of two inputs of the comparator circuit CMPp. An output signal ofthe triangular wave generator TWGEN is inputted to the other of the twoinputs of the comparator circuit CMPp. The oscillator OSC and thetriangular wave generator TWGEN are achieved by, for example, theconfiguration example of FIG. 25 referred to above.

<<Overall Circuit Operation of Power Supply Device [7]>>

FIG. 27 is a waveform diagram showing a schematic example of operationof FIG. 26. Firstly, when the oscillator OSC outputs a reset signal RT(‘H’ pulse), a duty control signal PWM is brought to an ‘H’ level (onlevel) due to the transition of the ‘H’ pulse to an ‘L’ level, so that atransistor Q1 is driven on. Thus, a growing input current Iin flowsthrough an inductor L1 via a transistor Q1, so that power is accumulatedin the inductor L1. Thereafter, when a set signal ST (‘H’ pulse) isoutputted from the comparator circuit CMPp, the duty control signal PWMis brought to an ‘L’ level (on level), so that the transistor Q1 isdriven off. In doing so, with the power accumulated in the inductor L1as an electromotive force, an output current Iout flows through a pathof a positive polarity output node Vout (+), a diode D1, and a negativepolarity output node Vout (−). Thus, a load circuit (not shown) isdriven by the output current Iout and an output capacitor Cout ischarged thereby.

When the transistor Q1 is on, the error amplifier circuit EA1 outputs anerror component of the output voltage Vout relative to the set voltage(Vr2). When the output voltage Vout is excessively high, the outputvoltage of the error amplifier circuit EA1 is lowered, so that a period(i.e., on duty of transistor Q1) from the output of a reset signal RT tothe output of the set signal ST from the comparator circuit CMPpdecreases. When the output voltage Vout is excessively low in reverse,the output voltage of the error amplifier circuit EA1 rises, so that theon duty of the transistor Q1 increases. Then, when the periodcorresponding to one cycle of the oscillator OSC has elapsed, thetransistor Q1 is driven on in response to the transition of the ‘H’pulse of the oscillator OSC to the ‘L’ level.

When such a control operation is used, for example, the peak value Ipkof the input current Iin flowing through the transistor Q1 whenreferring to FIG. 27 becomes Ipk=(Vin/K1)×Ton using the input voltageVin from a rectifying circuit DB1, the inductance value (L1) of theinductor L1, and the on time (Ton) of the transistor Q1. The averagevalue Iin_ave per one cycle (Tsw) of the duty control signal PWM isexpressed in the following equation (1D) using the peak value Ipk.

Iin_ave=(½)×(Ipk·Ton)/Tsw

=1/(2·L1)×(Ton² /Tsw)×Vin   (1D)

In the equation (1D), L1 and Tsw are fixed values, and Ton is a variableunaffected by Vin. For this reason, the input current Iin (ac currentIac that flows through AC power line) is placed in proportionality withVin and becomes a sinusoidal shape.

Eighth Embodiment

An eighth embodiment will explain one example in which a step-downconverter is configured by modifying the power supply topology of thehigh-side inverse converter described in the first embodiment.

<<Overall Circuit Configuration of Power Supply Device 8>>

FIG. 35 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to an eighth embodimentof the present invention. When a comparison is made between FIG. 35 andFIG. 1, the power supply device of FIG. 35 differs from the power supplydevice of FIG. 1 in that the reference node side at the output of therectifying circuit DB1 is coupled to the positive polarity output nodeVout (+) in the power supply device of FIG. 1, whereas in the powersupply device of FIG. 35, the reference node side of the rectifyingcircuit DB1 is coupled to the negative polarity output node Vout (−).That is, in FIG. 35, the rectifying circuit DB1 generates an inputvoltage Vin and an input current Iin at a node Ndb1 (a first node) onthe basis of the positive polarity output node Vout (−) (a second node).Coupling relations other than the above are similar to FIG. 1. The diodeD1 is coupled between a node Nsw (a third node) used as the source of atransistor Q1 and the negative polarity output node Vout (−) with theVout (−) side as its anode. A resistor Rcs for current detection iscoupled between the node Nsw (the third node) and one end (a fourthnode) of an inductor L1. The other end (a fifth node) of the inductor L1is coupled to the positive polarity output node Vout (+). A capacitorCout is coupled between the negative polarity output node Vout (−) andthe positive polarity output node Vout (+). Further, a load circuit LODlike light emitting diodes LEDs or the like is coupled in parallel tothe capacitor Cout.

<<Overall Circuit Operation of Power Supply Device [8]>>

In the power supply device of FIG. 35, firstly, current flows through apath of Ndb1, Q1, Rcs, L1, Vout (+) and Vout (−) when the transistor Q1is on. Hence, power is accumulated in the inductor L1 and an outputcurrent Iout is supplied to the load circuit LOD. On the other hand,when the transistor Q1 is off, current flows through a path of L1, Vout(+), Vout (−), D1 and Rcs with the power accumulated in the inductor L1as an electromotive force, so that an output current Iout is supplied tothe load circuit LOD. Even in such a configuration, the input currentIin (ac current Iac of commercial power source (AC) flows only during aperiod in which the transistor Q1 is on, in a manner similar to thepower supply device of FIG. 1. Since, however, the power supply deviceis equipped with the PFC circuit PIC1 including the square circuit SQdescribed above, a current waveform (Iac) close to the sinusoidal waveis obtained.

<<Advantages of Power Supply Device [8]>>

Thus, when the power supply device of FIG. 35 is used, an advantagesubstantially similar to that of the power supply device of FIG. 1 isobtained, and the following advantages are further obtained in additionto it. Firstly, an element low in breakdown voltage between the sourceand drain of the transistor Q1 can be used as the transistor Q1, and areduction in cost can hence be achieved. That is, while the source-drainvoltage Vds at the time of turning off of the transistor Q1 becomes“Vin+Vout” (Vout: voltage between the positive and negative polarityoutput nodes Vout (+) and Vout (−)), whereas in the power supply deviceof FIG. 35, the source-drain voltage Vds at the time of turning off ofthe transistor Q1 can be brought substantially to the input voltage Vin.Accordingly, an element lower in breakdown voltage than FIG. 1 by Vout(e.g., Vin: full-wave rectified value of 85 to 264 Vrms and Vout: 60 Vor 70 V or the like) can be used in FIG. 35.

Secondly, an improvement in power conversion efficiency can be achieved.Since the power supply devices shown in FIGS. 1 and 35 are both operatedin a current critical mode, they are both brought to zero currentswitching when the transistor Q1 is transitioned to on and hence enablesa reduction in switching loss as compared with a so-called hardswitching system. In the power supply devices shown in FIGS. 1 and 35,however, a difference occurs in the switching loss with theabove-described difference in Vds when the transistor Q1 is transitionedto off. FIGS. 36(a) and 36(b) are respectively typical diagrams eachshowing one example of the conditions that switching losses occur, inwhich FIG. 36(a) shows where the power supply device of FIG. 35 is used,and FIG. 36(b) shows, as its comparative example, where the power supplydevice of FIG. 1 is used. As shown in FIGS. 36(a) and 36(b), when thetransistor Q1 is transitioned to off, the falling waveform of a currentIm of the transistor Q1, and the rising waveform of the drain-sourcevoltage Vds of the transistor Q1 partly overlap each other. Therefore, aloss occurs in the overlapped portion (SLA1, SLA2).

When the power supply device of FIG. 35 is used at this time, the areaof the overlapped portion (SLA1) can be set smaller than that of theoverlapped portion (SLA2) in the power supply device of FIG. 1 becausethe value of the source-drain voltage Vds is small, thus enabling areduction in the switching loss. Described in more detail, a switchingloss Ps₁₃ off at the transition of the transistor Q1 to off is expressedin the following equation (10) assuming that Ip is taken as the peakvalue of the current Im. It is understood that reducing the value of thesource-drain voltage Vds upon reducing the switching loss is useful. Asa result of verification under Vin=100 Vrms, Vout=70 V and Pout (powerconsumption of load circuit LOD)=8 W by way of example, there could beobtained a result that the power conversion efficiency was 86.0% in thecase of the power supply device of FIG. 1, whereas the power conversionefficiency was 90.3% in the case of the power supply device of FIG. 35.

$\begin{matrix}{{P\; {s{\_ off}}} = {{\int_{0}^{t\; f}{{{(t)} \cdot {V(t)}}\ {{\int_{0}^{tf}\ {\left( {{I\; p} - {\frac{I\; p}{T\; f}t}} \right)\frac{V\; d\; s}{T\; f}t{t}}}}}} - {\frac{1}{6}{{Vds} \cdot I}\; {p \cdot t}\; f}}} & (10)\end{matrix}$

Incidentally, although not limited in particular, in FIG. 35, ametal-coated chip resistor having a resistance value such as 1.2 Ω, forexample can be used as the resistor Rcs for zero current detection withthe current critical mode in order to miniaturize an LED illuming deviceor the like. Its premise will be concretely explained. Firstly, if theload circuit LOD is of an LED of Vout=70 V, Pout=8 W and 100 Vrms, itscurrent consumption Iout becomes Iout=8 W/70 V=about 114 mA. In thiscase, a peak current Ip of an input current Iin at the peak of the inputvoltage Vin acts in the current critical mode and results in 2√{squareroot over (2)} times the current consumption Iout. If efficiency isassumed to be 90% and a power factor is assumed to be 0.9, then the peakcurrent Ip becomes Ip=0.114×2×√{square root over (2)}/(0.9×0.9)=398 mA.Now, assuming that Ip=0.5 A or the like in consideration of a margintaken upon determining the resistance value of the resistor Rcs, and theupper limit value of the detection voltage Vcs developed at the resistorRcs, which can be inputted to the PFC circuit PIC1, is 0.6 V, theresistance value of the resistor Rcs becomes Rcs=0.6 V/0.5 A=1.2Ω. Thus,when Rcs=1.2Ω or the like, power consumed or used up at the resistor Rcsis 0.114×0.114×1.2=15.6 mW, and a general resistor such as a ¼ W typecan be used.

Ninth Embodiment

A ninth embodiment will explain one example in which the configurationof a PFC circuit is modified using the power supply topology of thestep-down converter described in the eighth embodiment.

<<Overall Circuit Configuration of Power Supply Device [9]>>

FIG. 37 is a schematic diagram showing one example of a circuitconfiguration of a power supply device according to a ninth embodimentof the present invention. The power supply device shown in FIG. 37 isdifferent from the power supply device of FIG. 35 in terms of aninternal configuration of a PFC circuit and is similar thereto inconfiguration at other than it. In FIG. 37, the PFC circuit PIC11includes a ramp circuit RMP1, comparator circuits CMPp and CMPz, aset/reset latch circuit SRLT, a driver circuit DRV, and an erroramplifier circuit EA. The PFC circuit PIC11 of FIG. 37 is different fromthe PFC circuit PIC1 of FIG. 1 and serves as a critical converter whichis not equipped with a square circuit SQ and controls the on time usinga detection voltage Vcs developed from a resistor Rcs for currentdetection and a charging voltage by the ramp circuit RMP1.

The set/reset latch circuit SRLT drives a duty control signal PWM to an‘L’ level (off level) via the driver circuit DRV when a reset signal RTis outputted from the comparator circuit CMPp, and drives the dutycontrol signal PWM to an ‘H’ level (on level) via the driver circuit DRVwhen a set signal ST is outputted from the comparator circuit CMPz. Whenthe detection voltage Vcs developed by the resistor Rcs for currentdetection becomes lower than a predetermined comparison voltage Vr1, thecomparator circuit CMPz outputs the corresponding set signal ST. Whenthe voltage of the output signal Vs from the ramp circuit RMP1 exceedsan output voltage of the error amplifier circuit EA, the comparatorcircuit CMPp outputs the corresponding reset signal RT. The erroramplifier circuit EA amplifies a difference between the detectionvoltage Vcs and a comparison voltage Vr2 with the detection voltage Vcsas a negative polarity (−) input. The ramp circuit RMP1 includes acurrent source IS20, a capacitor C20 and a switch circuit SW20. When theduty control signal PWM is at the ‘L’ level (inverse output (/Q) of SRLTis at an ‘H’ level), the ramp circuit RMP1 fixes the output signal Vs toan ‘L’ level via the switch circuit SW20. When the duty control signalPWM is transitioned to the ‘H’ level, the ramp circuit RMP1 charges acurrent of the current source IS20 into the capacitor C20 to graduallyincrease the voltage of the output signal Vs.

<<Overall Circuit Operation of Power Supply Device [9]>>

FIG. 38 is a waveform diagram showing a schematic example of operationof the power supply device of FIG. 37. As shown in FIG. 38, when a setsignal ST is first inputted to the set/reset latch circuit SRLT, a dutycontrol signal PWM is transitioned to an ‘H’ level and thereby currentflows into the inductor L1 side via a transistor Q1. With its operation,a detection voltage Vcs developed at the resistor Rcs gradually rises.When the duty control signal PWM is transitioned to the ‘H’ level, theswitch circuit SW20 lying in the ramp circuit RMP1 is controlled to off,so that the voltage of an output signal Vs of the ramp circuit RMP1 alsogradually rises. On the other hand, the output voltage of the erroramplifier circuit EA is gradually lowered according to the rise in theVcs. Here, when the voltage of the output signal Vs reaches the outputvoltage of the error amplifier circuit EA, a reset signal RT isoutputted and correspondingly, the duty control signal PWM istransitioned to an ‘L’ level according to it. When the duty controlsignal PWM is transitioned to the ‘L’ level, a load circuit LOD isdriven by an electromotive force of the inductor L1, so that thedetection voltage Vcs is gradually lowered and the output voltage of theerror amplifier circuit EA gradually rises in reverse. When the dutycontrol signal PWM is transitioned to the ‘L’ level, the switch circuitSw20 lying inside the ramp circuit RMP1 is controlled to on, so that theoutput signal Vs is discharged to 0 V. Thereafter, when the detectionvoltage Vcs is lowered substantially to 0 V (when current IL1 flowingthrough the inductor L1 reaches zero), a set signal ST is inputted tothe set/reset latch circuit SRLT and similar operations are repeatedsubsequently.

<<Advantages of Power Supply Device [9]>>

Thus, when the power supply device of FIG. 37 is used, advantagessubstantially similar to FIG. 35 are obtained even though the powersupply device of FIG. 37 is lowered than that of FIG. 35 in powerfactor. Further, a size reduction in the power supply device and itscost reduction can be achieved as compared with the case of FIG. 35.That is, it is possible to eliminate the resistors Rac1 and Rac2 fordetection of the input voltage Vin in FIG. 35. It is however feared thatwith the reduction in the power factor, the present power supply deviceis not capable of satisfying a higher harmonic standard of illuminatingequipment, for example. According to the verification by the presentinventors or the like, however, there could be obtained a result thatthe standard could be sufficiently satisfied as shown in FIG. 39. FIG.39 is a diagram showing a result of verification of higher harmonicscontained in an input current Iin (ac current Iac flowing through an ACpower line) that flows in the power supply device of FIG. 37. Here, theverification is performed assuming that the input voltage Vin is 100Vrms, the output voltage Vout is 70 V and the power consumption Pout ofthe load circuit LOD is 8 W. Specification values of classes C eachindicative of the higher harmonic standard of the lighting equipment arealso shown in FIG. 39 along with it. As is understood from here, thestandard can be sufficiently satisfied even when the power supply deviceof FIG. 37 is used.

While the invention made above by the present inventors has beendescribed specifically based on the preferred embodiments, the presentinvention is not limited to the embodiments referred to above. It isneedless to say that various changes can be made thereto within thescope not departing from the gist thereof.

What is claimed is:
 1. An LED illuminating device comprising: a firstoutput node and a second output node coupled with a light emittingdiode; a rectifying circuit which rectifies ac power and supplies powerto a rectified output node; an N-channel type MOSFET having a gateterminal, a drain terminal and a source terminal, the drain terminalbeing coupled with the rectified output node; a current detectionresistor coupled with the source terminal and a ground node, the groundnode being coupled with a ground voltage; an inductor coupled with theground node and the first output node; a diode whose anode and cathodeare coupled with the second output node and the source terminal of theN-channel type MOSFET, respectively; an output capacitor coupled withthe first output node and the second output node; and a control circuitcoupled with the gate terminal and the source terminal of the N-channeltype MOSFET, a detection voltage of the source terminal being inputtedinto the control circuit, the control circuit providing a PWM signal tothe gate terminal to control the N-channel type MOSFET.